Semiconductor device

ABSTRACT

The present invention provides a semiconductor device realizing suppression of increase in consumption power. A semiconductor device has a signal line, a reception buffer circuit which is coupled to an end of the signal line and to which a signal is supplied from the signal line, and a delay element which is wired-OR coupled to an end of the signal line and shapes a waveform of a signal at the end of the signal line.

CROSS-REFERENCE TO RELATED APPLICATIONS

The disclosure of Japanese Patent Application No. 2016-030127 filed onFeb. 19, 2016 and No. 2015-223002 filed on Nov. 13, 2015 including thespecification, drawings and abstract is incorporated herein by referencein its entirety.

BACKGROUND

The present invention relates to a semiconductor device and relates to,for example, a semiconductor device having a signal line transmitting ahigh-speed signal and a semiconductor device having a plurality ofsemiconductor chips mounted over a print substrate and transmitting ahigh-speed signal among the semiconductor chips via signal lines of theprint substrate.

A semiconductor device capable of transmitting or receiving a high-speedsignal, for example, a signal whose transfer speed is 25 Gbps or higheris demanded. For example, in the case of transmitting/receiving signalsbetween semiconductor devices, the signals are transmitting/received byusing a serial communication system capable of reducing the influencecaused by a timing deviation among the signals received and alsosuppressing increase in the number of terminals. In this case, forexample, the semiconductor device is provided with a so-called SerDes(Serializer-Deserializer) circuit. A parallel signal formed by a processin the semiconductor device is converted to a high-speed signal (serialsignal) of 25 Gbps or higher by the SerDes circuit and the high-speedsignal is transmitted from the semiconductor device. The high-speedsignal of 25 Gbps or higher received by the semiconductor device isconverted to, for example, a parallel signal by the SerDes circuit, anda process on the parallel signal obtained by the conversion is performedin the semiconductor device.

The semiconductor device as described above is used for, for example,network control. In the case of using the semiconductor device fornetwork control, a semiconductor device having the SerDes circuit and asemiconductor device as a component of a volatile memory are mounted onan interposer which is mounted on a print substrate. The semiconductordevice having the SerDes circuit transmits/receives a high-speed signalof 25 Gbps or higher via a signal line formed in the print substrate.For example, the semiconductor device having the SerDes circuit controlsthe semiconductor device as a component of the volatile memory on thebasis of the received high-speed signal, writes data, converts data readfrom the semiconductor device as a component of the volatile memory to ahigh-speed signal (serial signal), and transmits the high-speed signalto another semiconductor device and/or an electronic device via thesignal line in the print substrate.

Since the high-speed signal is transmitted/received via the signal line,a distortion occurs in the signal waveform. To shape the distortedwaveform, for example, an analog filter circuit formed by a passiveelement was used. In recent years, to improve the reception performance,in place of the analog filter circuit, a digital filter circuit formedby an active element such as a transistor is used.

Patent literature 1 describes a technique related to a digital filtercircuit. Patent literature 2 describes a technique related to a signalline.

RELATED ART LITERATURE Patent Literature

-   Patent Literature 1: Japanese Unexamined Patent Application    Publication No. Hei 2 (1990)-72711-   Patent Literature 2: Japanese Unexamined Patent Application    Publication No. 2004-297411

SUMMARY

By using a digital filter circuit in place of an analog filter circuit,reception performance can be largely improved. However, when thetransmission speed of a signal increases to 32 Gbps and 54 Gbps,accordingly, the digital filter circuit has to be operated at clockfrequency of 32 GHz and 54 GHz.

The difficulty level of designing the digital filter circuit whichoperates at high clock frequency becomes high. Not only the difficultylevel of designing but also a problem that consumption power increasesdue to high-speed operation occurs.

Patent literature 1 discloses a technique related to a digital filtercircuit. It is assumed that the digital filter circuit described in thepatent literature 1 is formed by an active element. Consequently, in thecase of processing a high-speed signal, consumption power of the digitalfilter circuit increases. Patent literature 2 discloses a techniquerelated to a signal line. However, a technique of shaping deteriorationof a waveform which occurs due to propagation in a signal line is notdescribed.

The other problems and novel features will become apparent from thedescription of the specification and appended drawings.

To facilitate understanding of the problem, first, a technique examinedby the inventors of the present invention prior to the present inventionwill be described.

Examination by Inventors

FIGS. 21A and 21B are explanatory diagrams illustrating a techniqueexamined by the inventors of the present invention prior to the presentinvention. Illustrated in FIG. 21A are a signal line 2100, atransmission buffer circuit 2101, a digital filter circuit 2102, and areception buffer circuit 2103. From the transmission buffer circuit2101, signals are supplied in series to the signal line 2100. The signalfrom the transmission buffer circuit 2101 is supplied to the digitalfilter circuit 2102 (transmitted) via the signal line 2100, and anoutput from the digital filter circuit 2102 is supplied to the receptionbuffer circuit 2103.

The digital filter circuit 2102 and the reception buffer circuit 2103are included in an SerDes circuit provided for a first semiconductordevice and comprised of a plurality of active elements. The transmissionbuffer circuit 2101 is included in an SerDes circuit provided for asecond semiconductor device and comprised of a plurality of activeelements. An example of the active element is a transistor. The signalline 2100 is configured by a signal line formed on a print substrate onwhich the first and second semiconductor devices are mounted. With theconfiguration, a signal generated by the second semiconductor device isoutput from the transmission buffer circuit 2101 to the signal line2100, a signal from the signal line 2100 is supplied to the receptionbuffer circuit 2103 via the digital filter circuit 2102 in the firstsemiconductor device, and a signal output from the reception buffercircuit 2103 is processed in the first semiconductor device.

Since there is a loss in the signal line 2100, when a signal istransmitted in the signal line 2100, degradation occurs in the waveformof the signal (signal waveform).

In FIG. 21A, 2104 indicates a transmission waveform output from thetransmission buffer circuit 2101 to the signal line 2100. In FIG. 21A,2105 indicates a filter input waveform which is input from the signalline 2100 to the digital filter circuit 2102, and 2106 indicates afilter output waveform which is output from the digital filter circuit2102. For explanation, FIG. 21A illustrates the case where an impulsesignal whose voltage changes in an impulse state is output as thetransmission waveform 2104. In FIG. 21A, the horizontal axis of each ofthe transmission waveform 2104, the filter input waveform 2105, and thefilter output waveform 2106 indicates time, and the vertical axisindicates voltage.

When the transmission waveform 2104 whose voltage changes in an impulsestate is input to one of the ends of the signal line 2100, due to a lossin the signal line 2100, degradation occurs in the filter input waveformwhich is output from the other end of the signal line 2100. In theexample illustrated in FIG. 21A, the transmission waveform 2104 has animpulse-state waveform and, on the other hand, the filter input waveform2105 is a waveform of voltage which rises steeply and, after that,gradually decreases.

By properly setting the characteristics of the digital filter circuit2102, changes in the signal waveform in the signal line 2100 can beequalized, and the restored (shaped) signal waveforms can be output asthe filter output waveform 2106 from the digital filter circuit 2102. Bythe above, the filter output waveform 2106 can be made a waveformapproximated to the transmission waveform 2104. From the viewpoint ofequalization, the digital filter circuit 2102 can be regarded as anequalizer.

FIG. 21B is a block diagram illustrating the configuration of thedigital filter circuit 2102. The digital filter circuit 2102 hasaddition circuits SM1 and SM2, multiplication circuits MM1 to MMn, anddelay circuits DL1 to DLn. The addition circuits SM1 and SM2, themultiplication circuits MM1 to MMn, and the delay circuits DL1 to DLnare comprised of active elements such as transistors.

The addition circuit SM1 performs subtraction between the filter inputwaveform 2105 and the output of the addition circuit SM2, outputs theresult of the subtraction as the filter output waveform 2106, andsupplies it to the delay circuit DL1. The delay circuits DL1 to DLn arecoupled in series and each of delay circuits delays for predeterminedtime (delay time) and, after that, supplies the delayed signal to thedelay circuit in the next stage. Outputs of the delay circuits DL1 toDLn are supplied to corresponding multiplication circuits MM1 to MMn,respectively. The multiplication circuits MM1 to MMn performmultiplication between corresponding coefficients a₁ to a_(N) and thecorresponding delay circuits DL1 to DLn and supply the addition resultsto the addition circuit SM2. The addition circuit SM2 adds the outputsof the multiplication circuits MM1 to MMn and supplies the result ofaddition to the addition circuit SM1.

Delay time in each of the delay circuits DL1 to DLn corresponds to timeof one data width interval (UT delay). On the other hand, a signaloutput from the transmission buffer circuit 2101 to the signal line 2100changes in cycles according to the transmission speed of a signal whichis transmitted. The time of one data width interval is proportional tothe cycle of the transmission speed of the signal. Consequently, thevoltage of the filter output waveform 2106 which was output in the pastis multiplied with the coefficients a₁ to a_(N) and the resultant issubtracted from the filter input waveform 2105. Therefore, by settingproper coefficients a₁ to a_(N), the filter output waveform output fromthe digital filter circuit 2102 can be shaped to a proper waveform. InFIG. 21B, X_(k) indicates a digital value of the filter input waveform,and Y_(k) indicates a digital value of the filter output waveform.

To equalize degradations in the signal waveforms in the signal line2100, the digital filter circuit 2102 has a transfer function (inversetransfer function) inverse to the transfer function of the signal line2100. That is, it is set so that when the transfer function of thesignal line 2100 and the transfer function (inverse transfer function)of the digital filter circuit 2102 are multiplied, a constant isderived.

First, when the transfer function of the signal line 2100 is expressedin Z transformation notation, the transfer function S(z) of the signalline 2100 can be expressed as the equation (8) in FIG. 22. h_(k)indicates impulse response of the signal line 2100. Next, when thetransfer function H(z) of the digital filter circuit 2102 is expressedin Z transformation notation, it can be expressed as the equation (9) inFIG. 22, and the Z transformation code z can be expressed as theequation (10) in FIG. 22. UT expresses time in one data width interval.

When the transfer function S(z) of the signal line 2100 and the transferfunction H(z) of the digital filter circuit 2102 are multiplied, amultiplied transfer function H_(total) (z) is expressed as the equation(11) in FIG. 22. When the coefficient a_(k) is replaced as expressed bythe equation (12) in FIG. 22, the transfer function H_(total) (z) isexpressed as the equation (13) in FIG. 22. That is, the transferfunction H_(total) (z) becomes a constant, the waveform deteriorated bythe loss in the signal line 2100 is restored by the digital filtercircuit 2102, and the shaped filter output waveform 2106 can beobtained. The coefficient a_(k) corresponds to coefficients a₁ to a_(N)supplied to the multiplication circuits MM1 to MMn illustrated in FIG.21B. When a proper value of the coefficient a_(k) is obtained in someway, the signal waveform deteriorated through the signal line can berestored by numerical operation. In the equation (13), the multipliedtransfer function H_(total) (z) is an approximation equal sign andcoupled to a constant h_(o). To make is a complete equal sign (=), thenumber of multiplication circuits MM1 to MMn illustrated in FIG. 21B hasto be an infinite.

It is, however, impossible to make the number of multiplication circuitsMM1 to MMn infinite. By increasing the number of multiplicationcircuits, power consumption increases and the area occupied by thedigital filter circuit 2102 is also enlarged. Consequently, the numberof multiplication circuits MM1 to MMn is determined within the range ofthe allowable consumption power and the occupied area.

In the case of the digital filter circuit 2102, it is requested toselect any of the following two options. Option 1) The number ofarithmetic circuits and delay circuits configuring the digital filtercircuit are decreased to reduce consumption power. In the case ofselecting the option 1, waveform reproduction precision is suppressed.

Option 2) The number of arithmetic circuits and delay circuitsconfiguring the digital filter circuit are increased to make thewaveform reproduction precision high. In the case of selecting theoption 2, increase in power consumption and the occupied area occurs.That is, it is difficult to satisfy both reduction in consumption power(occupied area) and improvement of the waveform reproduction precision.Also in the case of selecting the option 1), the clock signal operatingthe arithmetic circuit and the delay circuit becomes high frequency, sothat the consumption power increases.

In the case of using the digital filter circuit 2102, there arerestrictions by not only the consumption power and occupied area butalso the sampling theorem. FIG. 23 is a diagram for explaining therestrictions by the sampling theorem. In the case of performing anumerical operation by the digital filter circuit 2102, the waveforms ofa signal have to be discretized every data width interval UT. That is,it is requested to sample the waveform in the sampling cyclecorresponding to the time of the one data width interval UT.

In FIG. 23, the filter input waveform 2105 is indicated in the upperleft part, the digital filter circuit 2102 and the reception buffercircuit 2103 are illustrated in the upper center part, and the filteroutput waveform 2106 is illustrated in the upper right part. In FIG. 23,the horizontal axis of each of the filter input waveform 2105 and thefilter output waveform 2106 indicates time, and the vertical axisindicates voltage. It is assumed here that the voltage of each of thefilter input waveform 2105 and the filter output waveform 2106 changesin the vertical directions with respect to predetermined voltage Vc as areference voltage.

In the lower center part in FIG. 23, the change in the voltage at theinput terminal of the reception buffer circuit 2103 is illustrated as aneye pattern (eye-diagram). The horizontal axis of the eye patternindicates time, and the vertical axis indicates voltage. The eye patternis generated by overlapping filter output waveforms supplied to theinput terminal of the reception buffer circuit 2103 when the same filterinput waveform is repeatedly supplied to the digital filter circuit2102.

FIG. 23 illustrates, for explanation, the case where the transferfunction H(z) of the digital filter circuit 2102 is 1. In each of thefilter input waveform 2105, the filter output waveform 2106, and the eyepattern, UT indicates time of one data width interval, and broken linesindicate sampling timings. The case where the sampling cycle as theinterval between the sampling timings is the same as the time of onedata width interval is illustrated.

In the digital filter circuit 2103, prior to the arithmetic process, thefilter input waveform 2105 is sampled at the sampling timings indicatedby the broken lines, and the arithmetic process is performed on discretedata obtained by the sampling. Consequently, even if the filter inputwaveform 2105 has a waveform close to a trapezoid shape, the filteroutput waveform after passing through the digital filter circuit 2102becomes a polygonal-line waveform as illustrated in FIG. 23. The eyepattern at the input terminal of the reception buffer circuit 2103becomes a rhomboid-shaped pattern obtained by overlapping the polygonalline waveforms. When the eye pattern becomes the rhomboid shape, if thetiming of fetching the voltage at the input terminal by the receptionbuffer circuit 2103 deviates, the reception level decreases. Forexample, in the eye pattern in FIG. 23, when the fetch timing deviatesfrom time t10 to t11, the reception level of the reception buffercircuit 2103 largely decreases.

By increasing the sampling frequency to, for example, several times andalso increasing the clock signal which operates the arithmetic circuitand the delay circuit configuring the digital filter circuit 2102 toseveral times, the filter output waveform 2106 closer to the filterinput waveform 2105 can be generated. Even if the fetch timing deviates,large decrease in the reception level can be prevented. However, sincethe frequency of the clock signal which operates the arithmetic circuitand the delay circuit configuring the digital filter circuit 2102becomes higher, the consumption power of the digital filter circuit 2102increases. In addition, when the clock frequency becomes higher, timingdesign of the delay circuit and the like becomes harder.

Means for Solving the Problems

A semiconductor device according to an embodiment has a signal line, afirst circuit which is coupled to an end of the signal line and to/fromwhich a signal is supplied from/to the signal line, and a delay elementwhich is wired-OR coupled to an end of the signal line and shapes thewaveform of a signal at the end of the signal line.

A part of a signal from the signal line or a signal to the signal lineis transmitted to the delay element at the wired-OR coupled end. In thedelay element, a returning wave based on the part of the signal suppliedis transmitted to the wired-OR coupled end. By the operation, thewaveform of the signal from the signal line or the signal to the signalline is shaped by the returning wave. Since the waveform of a signal isshaped by the delay element as a passive element, even the transmissionspeed of a signal passing through the signal line becomes higher,increase in consumption power at the time of shaping the waveform of thesignal can be suppressed. Therefore, the semiconductor device realizingsuppression of increase in consumption power can be provided.

According to the embodiment, the semiconductor device realizingsuppression of increase in consumption power can be provided.

BRIEF DESCRIPTION OF THE DRAWINGS

FIGS. 1A to 1C are diagrams illustrating a basic configuration of adigital filter according to a first embodiment.

FIGS. 2A to 2C are waveforms illustrating the operation of the digitalfilter according to the first embodiment.

FIG. 3 is a diagram for explaining the digital filter according to thefirst embodiment.

FIG. 4 is a block diagram illustrating a configuration that the digitalfilter according to the first embodiment is coupled to a transmissionbuffer circuit.

FIGS. 5A and 5B are tables comparing the digital filter according to thefirst embodiment and a digital filter circuit configured by an activeelement.

FIG. 6 is a cross section illustrating a section of a semiconductordevice according to the first embodiment.

FIG. 7 is a plan view illustrating a plane of a main part of thesemiconductor device according to the first embodiment.

FIG. 8 is a cross section illustrating an A-A′ section and a B-B′section of the plane depicted in FIG. 7.

FIGS. 9A and 9B are waveform charts illustrating waveforms of thedigital filter according to the first embodiment.

FIGS. 10A and 10B are waveform charts illustrating waveforms of thedigital filter according to the first embodiment.

FIG. 11 is a plan view illustrating a plane of a semiconductor deviceaccording to a modification of the first embodiment.

FIG. 12 is a cross section illustrating an A1-A1′ section and a B1-B1′section of the plane depicted in FIG. 11.

FIG. 13 is a plan view illustrating a plane in a semiconductor deviceaccording to a second embodiment.

FIG. 14 is a cross section illustrating an A2-A2′ section and a B2-B2′section of the plane depicted in FIG. 13.

FIG. 15 is a cross section illustrating a section of a semiconductordevice according to a third embodiment.

FIGS. 16A to 16C are a plan view and cross sections of the semiconductordevice according to the third embodiment.

FIG. 17 is a cross section illustrating a section of a semiconductordevice according to a fourth embodiment.

FIGS. 18A to 18C are a plan view and cross sections of the semiconductordevice according to the fourth embodiment.

FIG. 19 is a block diagram illustrating the configuration of a digitalfilter according to a fifth embodiment.

FIG. 20 is a block diagram illustrating the configuration of a digitalfilter according to a sixth embodiment.

FIGS. 21A and 21B are explanatory diagrams illustrating a techniqueexamined by the inventors of the present invention.

FIG. 22 is a diagram for explaining the technique examined by theinventors of the present invention.

FIG. 23 is a diagram for explaining the technique examined by theinventors of the present invention.

FIG. 24 is a waveform chart illustrating voltage waveforms of adifferential signal.

FIGS. 25A to 25C are diagrams illustrating eye patterns of a single-enddigital filter.

FIGS. 26A to 26C are diagrams illustrating eye patterns of a single-enddigital filter.

FIGS. 27A and 27B are diagrams illustrating the configuration of adigital filter according to a seventh embodiment and an equivalentcircuit.

FIGS. 28A to 28C are diagrams illustrating eye patterns according to theseventh embodiment.

FIGS. 29A to 29C are diagrams illustrating eye patterns according to theseventh embodiment.

FIG. 30 is a diagram for explaining the digital filter according to theseventh embodiment.

FIG. 31 is a plan view illustrating the structure of the digital filteraccording to the seventh embodiment.

FIG. 32 is a cross section illustrating the structure of the digitalfilter according to the seventh embodiment.

FIG. 33 is a plan view illustrating the structure of a digital filteraccording to a modification of the seventh embodiment.

FIG. 34 is a cross section illustrating the structure of the digitalfilter according to the modification of the seventh embodiment.

FIG. 35 is a plan view illustrating the structure of a digital filteraccording to an eighth embodiment.

FIG. 36 is a cross section illustrating the structure of the digitalfilter according to the eighth embodiment.

FIG. 37 is a plan view illustrating the structure of a digital filteraccording to a ninth embodiment.

FIG. 38 is a cross section illustrating the structure of the digitalfilter according to the ninth embodiment.

FIG. 39 is a plan view illustrating the structure of a digital filteraccording to a modification of the ninth embodiment.

FIG. 40 is a cross section illustrating the structure of the digitalfilter according to the modification of the ninth embodiment.

FIG. 41 is a block diagram illustrating the configuration of asemiconductor device according to a tenth embodiment.

FIG. 42 is a block diagram illustrating the configuration of asemiconductor device according to a modification of the tenthembodiment.

DETAILED DESCRIPTION

Hereinafter, embodiments of the present invention will be described indetail with reference to the drawings. In all of the drawings forexplaining the embodiments, in principle, the same reference numeralsare designated to the same parts and repetitive description will not begiven.

First Embodiment Basic Configuration of Digital Filter

First, the basic configuration of a digital filter provided in asemiconductor device according to a first embodiment will be described.The semiconductor device having therein the digital filter will bedescribed specifically later.

FIGS. 1A to 1C are diagrams illustrating a basic configuration of thedigital filter according to the first embodiment. FIG. 1A is a blockdiagram illustrating the configuration of the digital filter, FIG. 1B isan equivalent circuit diagram of the digital filter of FIG. 1A, and FIG.1C is a diagram of the transfer function of the digital filterillustrated in FIG. 1A.

In FIG. 1A, 1000 denotes a signal line (signal transmission path). It isassumed that the signal line 1000 has a pair of ends in FIG. 1A. In thediagram, SNO denotes one of ends of the signal line 1000 and SNI denotesthe other end of the signal line 1000. Illustrated in FIG. 1A are atransmission buffer circuit (second circuit) 1001, a digital filter1002, and a reception buffer circuit (first circuit) 1003.

As will be specifically described later, the semiconductor deviceaccording to the first embodiment has a print substrate, a plurality ofinterposers mounted on the print substrate, and semiconductor chipsmounted on the interposers. Since the semiconductor device has aplurality of semiconductor chips mounted on the print substrate, it canbe also regarded as an electronic device (including so-called SIP andMCM). However, in the specification, when it is unnecessary to clarify,a device including a print substrate, interposers, and semiconductorchips will be also called a semiconductor device. Similarly, in thepresent specification, when it is unnecessary to clarify, a devicehaving interposers and semiconductor chips mounted on the interposerswill be also called a semiconductor device. Further, in thespecification, when it is unnecessary to clarify, a semiconductor chipwill be also called a semiconductor device.

As will be specifically described later, the reception buffer circuit1003 illustrated in FIG. 1A is formed in a first semiconductor chipmounted on a first interposer, and the transmission buffer circuit 1001is formed in a second chip mounted on a second interposer different fromthe first interposer. The first and second interposers are mounted onthe same print substrate. The signal line 1000 illustrated in FIG. 1Aexpresses a signal wire electrically coupling the transmission buffercircuit 1001 formed in the second semiconductor chip and the receptionbuffer circuit 1003 formed in the first semiconductor chip. The signalwire includes, for example, a signal wire (wiring pattern) formed in theprint substrate.

The digital filter 1002 has a delay element DLN having a pair of endsDN1 and DN2. The delay element DLN is comprised of a delay wire (signalwire) having a predetermined length, and a pair of ends of the signalwire corresponds to the pair of ends DN1 and DN2. The end DN2 of thedelay element DLN is wired-OR-coupled to an end SNO of the signal line1000. Specifically, at a node WRN, the end DN2 of the delay element DLNand the end SNO of the signal line 1000 are electrically coupled. Thenode WRN is electrically coupled to the input terminal (input node) ofthe reception buffer circuit 1003. The other end DN1 of the delayelement DLN is electrically coupled to predetermined voltage Vs. In FIG.1A, the predetermined voltage Vs is ground voltage of the circuit.

The transmission buffer circuit 1001 receives a serial signal to betransmitted and supplies it to an end SNI as one of ends of the signalline 1000. The supplied serial signal passes through the signal line1000 and reaches the end SNO of the signal line 1000. The signal reachedthe end SNO of the signal line 1000 is distributed to the receptionbuffer circuit 1003 and the digital filter 1002. The ratio ofdistribution is indicated as a distribution ratio “b” of the signal. Thesignal of the distribution ratio “b” in signals at the end SNO of thesignal line 1000 is input (supplied) as an input signal FW to the endDN2 of the delay element DLN. The remaining signal, that is, a signal1-b is supplied to the input terminal of the reception buffer circuit1003.

Since the other end DN1 of the delay element DLN in the digital filter1002 is coupled to the predetermined voltage Vs (ground voltage of thecircuit), the impedance of the other end DN1 of the delay element DLN issmaller than that of the end DN2 of the delay element DLN. Consequently,the input signal FW input to the end DN2 of the delay element DLNreturns on the side of the other end DN1, and a returning signalindicated by a broken line is output as an output signal RW from the endDN2 of the delay element DLN to the wired-OR-coupling part (node WRN).Since the delay element DLN has a loss, the output signal RW output fromthe end DN2 attenuates as compared with the input signal FW input to theend DN2. The output signal RW is delayed from the input signal FW. Sincethe end DN2 of the delay element DLN which is wired-OR-coupled in thenode WRN is an end at which a signal is input and output, the end DN2can be regarded as an input/output end or an input/output terminal.

FIG. 1B is an equivalent circuit diagram of the delay element DLNillustrated in FIG. 1A. The delay element DLN is expressed by adistributed constant circuit. Although not limited, each of a pluralityof distributed constant circuits is expressed as a π-type distributedconstant circuit, and the π-type distributed constant circuit iscomprised of an inductance L, a resistance R, a capacitance C, and aconductance G. The equivalent circuit of the delay element DLN isexpressed so that a plurality of inductances L and a plurality ofresistors R are coupled in series between the ends DN2 and DN1, and aplurality of conductances G and capacitances C are coupled in parallelbetween the delay element DLN and the predetermined voltage Vs (theground voltage of the circuit).

In the equivalent circuit illustrated in FIG. 1B, as described above,the delay element DLN has a loss due to the influence of the inductanceL, the resistance R, the capacitance C and so on, and the output signalRW attenuates as compared with the input signal FW. When the attenuationcoefficient of the signal (signal attenuation coefficient) is set asβ/2, the signal attenuation coefficient of a period in which a signal isinput to the end DN2 is output from the end DN2 (round-trip signalattenuation coefficient) in the delay element DLN is expressed ase^(−β). Since a round trip of a signal is considered, the attenuationcoefficient of a round trip is expressed as β/2λ2=β. On the other hand,delay time required for a signal to make a round trip (round-trip delaytime) is determined by the ratio UT/m between the data width interval UTand a coefficient “m”. In this case, the coefficient “m” is an integer1, 2, 3, . . . . In FIG. 1A, the signal making a round trip in the delayelement DLN is drawn as the input signal FW and the output signal(returning wave) RW.

The output signal RW output from the end DN2 of the delay element DLN iscombined with a signal from the signal line 1000 at the node WRN by thewired-OR coupling. Since the output signal RW is a returning wave in thecombination, it works so as to decrease the absolute value of the signalfrom the signal line 1000. Since the round-trip delay time of the signalin the delay element DLN is a fraction of an integer of one data widthinterval UT, the absolute value of the signal from the signal line 1000is adjusted so as to decrease by one or plural returning waves from thedelay element DLN. If the round-trip delay time is not a fraction of theinteger of the one data width interval UT, it is considered that thereturning wave from the delay element DLN works on the signal from thesignal line 1000 so as to increase the absolute value at the node WRNand the signal from the signal line 1000 deteriorates.

When the end DN1 of the delay element DLN is in the floating state, theimpedance in the end DN1 becomes higher than that in the end DN2, andthe output signal RW output from the end DN2 of the delay element DLNcomes to have a traveling wave. As a result, it may happen that thetraveling wave is combined with the signal from the signal line 1000 inthe node WRN and a signal supplied to the input terminal of thereception buffer circuit 1003 becomes a deteriorated signal.Consequently, in the first embodiment, the end DN1 of the delay elementDLN is coupled to the predetermined voltage Vs.

The time of the round-trip signal delay UT/m is determined so as to be afraction of an integer of the one data width interval UT inconsideration of the length of the delay element DLN, that is, thedistance between the ends DN1 and DN2 and width, thickness, material,and the like of the delay wire configuring the delay element DLN.

FIG. 1C is a diagram illustrating the transfer function of the delayelement DLN. In FIG. 1C, the equation (1) expresses the transferfunction H(z) of the delay element DLN in Z transformation notation. Inthe equation (1), s indicates Laplace coefficient. As described above,“b” indicates the distribution ratio of the signal, UT indicates onedata width interval, and m expresses an integer.

Operation of Digital Filter

Next, the operation of the digital filter 1002 illustrated in FIG. 1Awill be described. FIGS. 2A to 2C are waveform charts illustrating theoperation of the digital filter 1002. The horizontal axis of each of thediagrams indicates time, and the vertical axis indicates voltage. FIG.2A indicates the transmission waveform 2104 supplied from thetransmission buffer circuit 1001 (FIG. 1A) to the end SNI of the signalline 1000 (FIG. 1A). FIGS. 2B and 2C indicate the waveforms of thesignal in the part (node WRN) which is wired-OR coupled. FIG. 2Bindicates combination of the filter input waveform 2105 from the signalline 1000 and the waveform of the output signal RW from the delayelement DLN realized by the wired-OR coupling in the node WRN. FIG. 2Cindicates the filter output waveform 2106 formed by the coupling in FIG.2B. Since the signal at the node WRN is supplied to the input terminalof the reception buffer circuit 1003, it can be regarded that FIG. 2Cillustrates the waveform of the input signal or reception signal of thereception buffer circuit 1003.

A serial signal according to a predetermined transmission speed issupplied to the transmission buffer circuit 1001, and a transmissionsignal corresponding to the supplied serial signal is supplied to theend SNI of the signal line 1000. To facilitate explanation, thetransmission waveform 2104 whose voltage changes in an impulse state isinput to the end SNI of the signal line 1000.

The transmission wavelength 2104 input to the end SNI of the signal line1000 is transmitted to the end SNO of the signal line 1000. Since thesignal line 1000 has a loss, a waveform deteriorated from thetransmission waveform 2104 is generated as the filter input wavelength2105. Since the transmission waveform 2104 changes in an impulse shape,as illustrated in FIG. 2B, the filter input waveform 2105 output fromthe transmission line 1000 has a shape that the voltage rises steeplyand gradually decreases. A part of the output signal (filter inputwaveform 2105) from the signal line 1000 is input to the end DN2 of thedelay element DLN in the node WRN. As described with reference to FIG.1A, a part of the supplied output signal (filter input waveform 2105) isoutput to the node WNR as an attenuated returning wave after theround-trip delay time in the delay element DLN.

That is, after the round-trip delay time, the output signal RW from thedelay element DLN is transmitted to the part of the wired-OR-coupling.Since the waveform at this time is a returning wave, it is a waveformwhose phase is inverted from that of the output signal (filter inputwaveform 2105), and the value of the output signal is a value attenuatedfrom the output signal (filter input waveform 2105). In thewired-OR-coupling part, that is, the node WNR, the output signal (filterinput waveform 2105) and the output signal RW (waveform of the returningwave) from the delay element DLN are combined. In FIG. 2B, the outputsignal RW generated by a round trip in the delay element is indicated bythe reference numeral (1). The round trip is generated infinite numberof times. In FIG. 2B, as an example, the output signals RW generated bysecond to sixth round trips are indicated by reference numerals (2) to(6). As attenuation occurs each time a signal makes a round trip in thedelay element DLN, the value of the output signal RW generated by around trip gradually decreases.

It is desirable to provide the wired-OR-coupling part, that is, the nodeWNR near the input terminal of the reception buffer circuit 1003. Thereason is as follows. An impedance exists also between the node WNR andthe input terminal of the reception buffer circuit 1003. When the nodeWNR and the input terminal of the reception buffer circuit 1003 areapart, the impedance increases and it becomes undesirable when the valueof the impedance is considered. The resistance R or conductance G perunit length of the delay element DLN is larger than the resistance orconductance per unit length of the signal line 1000.

Since the filter input waveform 2105 at the node NRN and the waveform ofthe output signal RW (for example, the waveforms of the referencenumerals (1) to (6)) are combined by wired-OR-coupling, the waveformsare overlapped. As a result, the filter output waveform 2106 supplied tothe input terminal of the reception buffer circuit 1003 has a shapesimilar to the transmission waveform 2104 as illustrated in FIG. 2C. Asa result, the filter output waveform 2106 supplied to the input terminalof the reception buffer circuit 1003 comes to have a shape similar tothe transmission waveform 2104 as illustrated in FIG. 2C. That is, thewaveform is restored (shaped).

The operation of the delay element DLN will be described more briefly asfollows. A part of a filter input signal (filter input waveform 2105)sent from the signal line 1000 is input to the end DN2 of the delayelement DLN. However, as there is no exit, the part returns to the endDN2 (input terminal). At this time, the resistance or conductance perunit length of the delay element DLN is made larger than that of thesignal line 1000, and the end DN1 of the delay element DLN is pinned tothe predetermined voltage Vs such as the ground voltage of the circuit.By the operation, the phase as illustrated by the reference numeral (1)in FIG. 2B (the polarity using the predetermined voltage Vs as areference) is inverted, and the attenuated signal returns to thewired-OR-coupling part (node WRN). Since the impedance of thewired-OR-coupling part (node WRN) and that of the end DN1 of the delayelement DLN are different, the signal once input from the signal line1000 to the delay element DLN repeats returning in the delay elementDLN, the output signal RW attenuated indicated by the reference numerals(2) to (6) or the like returns to the wired-OR-coupling part and iscombined with the filter input waveform 2105.

The filter input waveform 2105 passed though the signal line 1000 has ashape which is trailed from the impulse-shaped waveform (rectangularwaveform) as illustrated in FIG. 2B and is combined with the group ofwaveforms (waveforms indicated by the reference numerals (1) to (6) andthe like) of the output signal RW generated by the delay element DLN,thereby eliminating the tail part to reconstruct a signal waveform closeto the original rectangular wave.

In FIG. 2B, UT indicates one data width interval as described above.Time tsa indicates the timings at which the output signal RW from thedelay element DLN becomes a peak. Consequently, it can be regarded thattime between adjacent times tsa corresponds to round-trip delay time inthe delay element DLN. Since FIGS. 2A to 2C illustrate the case wherethe coefficient “m” described with reference to FIGS. 1A to 1C is 1, thetime between the adjacent times tsa is the same as the time of one datawidth interval UT.

By increasing the coefficient “m” described with reference to FIGS. 1Ato 1C in integers, the number of round trips in the delay element DLN inpredetermined time can be increased. That is, the number of waveforms ofthe output signal RW of the delay element DLN combined with the filterinput waveform 2105 can be increased in predetermined time, and thefilter output waveform 2106 supplied to the input terminal of thereception buffer circuit 1003 can be made closer to the transmissionwaveform 2104. The coefficient “m” is preferably, for example, aboutfour.

As described with reference to FIGS. 2A to 2C, in the case of performinga process by the digital filter circuit 2102, the filter input waveform2105 from the signal line 2100 is sampled and a process is performed bythe arithmetic circuit. In this case, the times tsa illustrated in FIG.2B can be regarded as sampling timings of sampling the filter inputwaveform 2105, and an arithmetic operation by the arithmetic circuit isperformed on digital values obtained by the sampling using the timebetween the adjacent times tsa as the sampling cycle.

In the first embodiment, the digital filter is configured by the delayelement DLN formed by the delay wire as the passive element.Consequently, a signal waveform in which deterioration occurs due totransmission through the signal line 1000 can be restored (shaped) whilereducing power consumption. When the time tsa illustrated in FIG. 2B isconsidered as a sampling timing, the digital filter according to thefirst embodiment can be regarded as equivalently infinite number ofarithmetic circuits in arbitrary sampling cycles. While reducing theconsumption power, the signal waveform can be restored (shaped) withhigh precision.

Although not illustrated in FIG. 1A, each of the signal line 1000 andthe delay element DLN is disposed so as to be parallel to a voltage wireto which predetermined voltage is supplied. The end DN1 of the delayelement DLN is coupled to the voltage wire disposed in parallel to thedelay element DLN. A signal loss per unit length in the delay elementDLN and a voltage wire disposed in parallel to the delay element DLN isset larger than that per unit length of the signal line.

Next, the digital filter 1002 illustrated in FIG. 1A will be describedusing the transfer function in the Z transformation notation. Asillustrated in FIG. 21B, the digital filter circuit 2102 is comprised ofan active element. On the other hand, as illustrated in FIG. 1A, thedigital filter 1002 according to the first embodiment is comprised of apassive element. Also even in the filter comprised of the passiveelement, digital computation is performed on the transfer function aswill be described later. Consequently, in the present specification,although the filter is comprised of the passive element, it is called adigital filter.

FIG. 3 is a diagram for explaining the digital filter according to thefirst embodiment. In a manner similar to the above description, theround-trip delay time of the delay element DLN is expressed as UT/m. Inthis case, the coefficient “m” is an integer 1, 2, 3, 4 . . . . Theround-trip signal attenuation rate of the delay element DLN is expressedas e^(−β).

The transfer function H(z) of the digital filter 1002 is expressed bythe equation (2) in FIG. 3. In this case, “b” indicates the distributionratio of the signal and b₀, c, and γ indicate variables. On the otherhand, the transfer function S(z) of the signal line 1000 is expressed bythe equation (3) in FIG. 3. In the equation (3), ho and “a” indicatevariables. The deterioration factors of deteriorating signalspropagating through the signal line 1000 are a skin effect and adielectric loss. In the equation (3), α₁ indicates a loss by the skineffect and α₂ indicates the dielectric loss.

The total transfer function H_(total)(z) obtained by multiplying thetransfer function of the signal line 1000 and the transfer function ofthe digital filter 1002 is expressed by the equation (4) in FIG. 3. Fromthe equation (4), a condition that the transfer function H_(total) (Z)becomes a constant ho which is substantially constant as expressed bythe equation (5) in FIG. 3 exists. For example, by making a setting asexpressed by the equation (6) in FIG. 3, the constant ho which issubstantially constant is obtained. That is, by using the digital filter1002, the signal line 1000 can be equalized.

In the equation (4), H(z)S(z) expresses the case where the digitalfilter 1002 is provided on the side of the transmission buffer circuit1001, and S(z)H(z) expresses the case where the digital filter 1002 isprovided on the side of the reception buffer circuit 1003. That is, thedigital filter 1002 may be coupled near the input terminal of thereception buffer circuit 1003 or near the output terminal of thetransmission buffer circuit 1001.

FIG. 4 is a block diagram illustrating a configuration in the case ofcoupling the digital filter 1002 to the output terminal of thetransmission buffer circuit 1001. FIG. 4 is similar to FIG. 1A and thedifferent point is that the digital filter 1002 is coupled to the outputterminal of the transmission buffer circuit 1001. Specifically, apredetermined part of a signal wire coupling the output terminal of thetransmission buffer circuit 1001 and the end SNI of the signal line 1000is the node WRN, and the end DN2 of the delay element DLN as a componentof the digital filter 1002 is wired-OR-coupled to the node WRN.

In this case, the waveform of a signal in the node WRN is deformed(adjusted) in advance by an output signal (returning wave) from thedigital filter 1002 so that the waveform of a signal output from the endSNO of the signal line 1000 is shaped to a waveform similar to thetransmission waveform output from the output terminal of thetransmission buffer circuit 1001. Since the operation of the digitalfilter 1002 is similar to that described with reference to FIGS. 1A to1C to FIG. 3, the description will not be repeated.

FIGS. 5A and 5B are tables of comparison between the digital filter 2102and the digital filter 1002. FIG. 5A illustrates comparison in thefunctional blocks and FIG. 5B illustrates comparison in functions.

As illustrated in FIG. 21B, the digital filter circuit 2102 is comprisedof function blocks such as the multiplication circuits MM1 to MMn, theaddition circuits SM1 and SM2, and the delay circuits DL1 to DLn. In thedigital filter 1002 according to the first embodiment, the functionblocks are replaced by physical coupling of physical amounts and theends of the delay element DLN. Specifically, as illustrated in FIG. 5A,the “n-th multiplication circuit” in the digital filter circuit 2102 isreplaced by “loss in delay element after n times of round trips” in thedigital filter 1002, and the “n-th delay circuit” is replaced by “delayin delay element after n times of round trips”. Further, “additioncircuit” in the digital filter circuit 2102 is replaced by“wired-OR-coupling to signal line and predetermined voltage coupling(phase inversion)” in the digital filter 1002.

FIG. 5B illustrates the case of comparing the functions between thedigital filter circuit 2102 and the digital filter 1002 according to thefirst embodiment. Specifically, in the digital filter circuit 2102, only“finite number” of multiplication circuits can be provided in reality.In contrast, in the digital filter 1002, the function of amultiplication circuit is realized by a loss of the delay element, sothat “infinite number” of equivalent multiplication circuits can beprovided. Similarly, in the digital filter circuit 2102, only “finitenumber” of delay circuits can be provided in reality. In contrast, inthe digital filter 1002, the function of a delay circuit is realized bya delay of the delay element DLN, so that “infinite number” ofequivalent delay circuits can be provided.

Further, in a sampling cycle of sampling a signal transmitted in thesignal line is one data width interval “UT” in the digital filtercircuit 2012 and, in contrast, an equivalent sampling cycle in thedigital filter 1002 is “arbitrary”. The equivalent sampling cyclecorresponds to the round-trip delay time UT/m. Although the equivalentsampling cycle is “arbitrary”, desirably, it is set to round-trip delaytime (equivalent sampling cycle) determined by the coefficient “m” of aninteger.

The degree of freedom of multiplier coefficients a₁ to a_(N) (FIG. 21)supplied to the multiplication circuits configuring the digital filtercircuit 2102 can be “arbitrarily” determined in the digital filtercircuit 2102. In the digital filter 1002, as an equivalent multipliercoefficient is realized by attenuation of the delay element DLN, thedegree of freedom of the multiplier coefficient is “only monotonicdecrease”. However, for example, by amplifying a signal supplied to theinput terminal of the reception buffer circuit 1003, a signal at anarbitrary voltage level can be output from the reception buffer circuit1003.

In the digital filter 1002 according to the first embodiment, themultiplier coefficient is “only monotonic decrease” as written in FIG.5B. It is consequently desirable to amplify the waveform of areconstructed (shaped) signal in the reception buffer circuit 1003 andthe like.

As described above, in the digital filter 1002 according to the firstembodiment, in a manner similar to the digital filter circuit 2102illustrated in FIG. 21, the signal line 1000 can be equalized, andconsumption power can be reduced. The number of equivalentmultiplication circuits is infinite and time resolution of the digitalfilter can be made many times of that of the digital filter circuit2102. Therefore, large reduction of consumption power and improvement insignal waveform decoding precision can be realized.

General Configuration of Semiconductor Device

Next, the general configuration of the semiconductor device according tothe first embodiment will be described. In the semiconductor devicedescribed here, the digital filter 1002 described in FIG. 1 isincorporated. FIG. 6 is a cross section illustrating a section of asemiconductor device 6000 according to the first embodiment. Thesemiconductor device 6000 has a print substrate PBS, a plurality ofpackage substrates mounted over the print substrate PBS, and a pluralityof interposers mounted over the package substrates. Further, over eachof the interposers, a semiconductor chip is mounted. Consequently, thesemiconductor device 6000 according to the first embodiment can beregarded as an electronic device having a plurality of semiconductorchips.

To facilitate description, in FIG. 6, two package substrates PPS-1 andPPS-2 mounted over the print substrate PBS are illustrated. Theinterposer mounted over the package substrate PPS-1 is indicated asreference numeral INS-1, and the interposer mounted over the packagesubstrate PPS-2 is indicated as reference numeral INS-2.

FIG. 6 illustrates the case where, although not limited, a plurality ofsemiconductor chips MCH-1 to MCH-4 (third semiconductor chips) and LCH-1(first semiconductor chip) are mounted over the first interposer INS-1(first interposer), and one semiconductor chip LCH-2 (secondsemiconductor chip) is mounted over the interposer INS-2 (secondinterposer). Obviously, the number of semiconductor chips mounted overeach of the interposers and the like is an example and the presentinvention is not limited to the number. In the semiconductor device 6000according to the first embodiment, the semiconductor chips MCH-1 toMCH-4 as a part of the semiconductor chips mounted over the interposerINS-1 are stacked stereoscopically (three-dimensionally) and thesemiconductor chip LCH-1 as a part of the semiconductor chips is mountedflatly (two-dimensionally). Specifically, in the top view of theinterposer INS-1, the semiconductor chips MCH-1 to MCH-4 are disposed soas to overlap one another in a predetermined first region in theinterposer INS-1, and the semiconductor chip LCH-1 is disposed in apredetermined second region different from the predetermined firstregion in which the semiconductor chips MCH-1 to MCH-4 are disposed.

Each of the semiconductor chips MCH-1 to MCH-4 is, for example, a memorysemiconductor chip storing information, and the semiconductor chip LCH-1is a logic semiconductor chip performing transmission/reception ofinformation, control, and the like on the memory semiconductor chipsMCH-1 to MCH-4. The logic semiconductor chip LCH-1 can be alsostereoscopically stacked over the interposer INS-1. For example,although the memory semiconductor chips MCH-1 to MCH-4 may be stackedover the logic semiconductor chip LCH-1, since each of the semiconductorchips generates heat, as illustrated in FIG. 6, it is desirably tostereoscopically stack only the memory semiconductor chips MCH-1 toMCH-4 and mount the logic semiconductor chip in the second regiondifferent from the memory semiconductor chips MCH-1 to MCH-4.

The print substrate PBS has a first main surface PBF1 and a second mainsurface PBF2 opposed to the first main surface PBF1 and further has,between the first and second main surfaces PBF1 and PBF2, a plurality ofconductive layers and a plurality of insulating layers alternatelysandwiched. The plurality of conductive layers are stacked whilesandwiching the insulating layers so as to be electrically separated. Onthe first main surface PBF1 of the print substrate PBS, a plurality ofball electrodes PDE are formed. In FIG. 6, to avoid the drawing frombecoming complicated, the reference characters PDE are designated onlyto the rightmost ball electrode and the leftmost ball electrode. Theball electrodes formed over the first main surface PBF1 of the printsubstrate PBS, for example, two ball electrodes are electrically coupledto each other via a predetermined wiring pattern formed by theconductive layer in the print substrate PBS. In FIG. 6, as an example ofwiring patterns formed by the conductive layers in the print substratePBS, wiring patterns (signal wires) PBL1 and PBL2 are illustrated.

The package substrate PPS-1 also has a first main surface PPF1 and asecond main surface PPF2 opposed to the first main surface PPF1 and hasa plurality of conductive layers and a plurality of insulating layerssandwiched between the first and second main surfaces PPF1 and PPF2. Theplurality of conductive layers are stacked while sandwiching theinsulating layers. On the first main surface PPF1 of the packagesubstrate PPS-1, a plurality of bump electrodes (not illustrated) areformed. On the second main surface PPF2 of the package substrate PPS-1,a plurality of ball electrodes (not illustrated) are formed. The densityof the bump electrodes formed on the first main surface PPF1 is set tobe higher than that of the ball electrodes formed on the second mainsurface PPF2. A desired wiring pattern is formed by the conductivelayers sandwiched between the first and second main surfaces PPF1 andPPF2. By the formed wiring pattern, for example, the bump electrodeformed on the first main surface PPF1 and the ball electrode formed onthe second main surface PPF2 are electrically coupled. FIG. 6illustrates an example of the wiring patterns formed by the conductivelayers sandwiched between the first and second main surfaces PPF1 andPPF2, and a part of the patterns are drawn as wiring patterns (signalwires) PPL1-1 and PPL1-2.

The interposer INS-1 also has a first main surface INF1 and a secondmain surface INF2 opposed to the first main surface INF1 and has aplurality of conductive layers and a plurality of insulating layerssandwiched between the first and second main surfaces INF1 and INF2. Theplurality of conductive layers are stacked while sandwiching theinsulating layers. On the first main surface INF1 of the interposerINS-1, a plurality of microbump electrodes (not illustrated) are formed.On the second main surface INF2 of the interposer INS-1, a plurality ofbump electrodes (not illustrated) are formed. The density of themicrobump electrodes formed on the first main surface INF1 is set to behigher than that of the bump electrodes formed on the second mainsurface INF2.

Desired wiring patterns are formed by the conductive layers sandwichedbetween the first and second main surfaces INF1 and INF2. By a formeddesired first wiring pattern, for example, the microbump electrodeformed on the first main surface INF1 and the bump electrode formed onthe second main surface INF2 are electrically coupled. By a formeddesired second wiring pattern, the microbump electrodes formed on thefirst main surface INF1 are electrically coupled. FIG. 6 illustrates anexample of the first wiring pattern electrically coupling the microbumpelectrode and the bump electrode and the second wiring patternelectrically coupling the microbump electrodes. The first wiringpatterns each coupling a predetermined microbump electrode and a bumpelectrode in the first wiring patterns are drawn as wiring patterns(signal wires) INL1-1 and INL1-2.

A plurality of microbump electrodes (not illustrated) are formed overthe main surface of the memory semiconductor chip MCH-1, each of thememory semiconductor chips MCH-2 to MCH-4 has a plurality of electrodes(not illustrated) in its main surface, and each of the electrodes of thememory semiconductor chips MCH-2 to MCH-4 is electrically coupled to themicrobump electrode of the memory semiconductor chip MCH-1 via a throughhole THF. Consequently, an internal circuit block in each of the memorysemiconductor chips MCH-1 to MCH-4 is electrically coupled to themicrobump electrode of the memory semiconductor chip MCH-1.

The logic semiconductor chip LCH-1 has a first main surface CHF1 and asecond main surface CHF2 opposed to the first main surface CHF1. Thelogic semiconductor chip LCH-1 has an SerDes circuit. Although theSerDes has a plurality of transmission buffer circuits and a pluralityof reception buffer circuits, in FIG. 6, one transmission buffer circuitis illustrated with reference numerals SCB1-1 and two reception buffercircuits are illustrated with reference numerals RCB1-1 and RCB1-2. Overthe second main surface CHF2 of the logic semiconductor chip LCH-1, aplurality of microbump electrodes (not illustrated) are formed. To themicrobump electrodes, a circuit block in the logic semiconductor chipLCH-1 is electrically coupled. FIG. 6 illustrates a state where onetransmission buffer circuit SCB1-1 and two reception buffer circuitsRCB1-1 and RCB1-2 are coupled to the microbump electrodes.

The memory semiconductor chip MCH-1 and the logic semiconductor chipLCH-1 are mounted so that the main surface of the memory semiconductorchip MCH-1 and the second main surface CHF2 of the interposer INS-1 areopposed to the first main surface INF1 of the interposer INS-1. At thistime, the memory semiconductor chips MCH-2 to MCH-4 are mounted so as tobe stacked stereoscopically over the memory semiconductor chip MCH-1.

The microbump electrodes formed over the main surface of the memorysemiconductor chip MCH-1 and the microbump electrodes formed over thesecond main surface CHF2 of the logic semiconductor chip LCH-1 areelectrically coupled to the microbump electrodes formed over the firstmain surface INF1 of the opposed interposer INS-1 by microbumps MBM.

The interposer INS-1 is mounted over the package substrate PPS-1 so thatthe second main surface INF2 is opposed to the first main surface PPF1of the package substrate PPS-1. At this time, the bump electrodes formedover the first main surface PPF1 of the package substrate PPS-1 and thebump electrodes formed over the second main surface INF2 of theinterposer INS-1 are electrically coupled by bumps SBM. The packagesubstrate PPS-1 is mounted over the print substrate PBS so that thesecond main surface PPF2 is opposed to the first main surface PBF1 ofthe print substrate PBS. At this time, the ball electrodes formed overthe second main surface PPF2 of the package substrate PPS-1 and the ballelectrodes formed over the first main surface PBF1 of the printsubstrate PBS are electrically coupled by balls SBL.

By the plurality of microbumps, the memory semiconductor chips MCH-1 toMCH-4 and the logic semiconductor chip LCH-1 are coupled tocorresponding microbump electrodes in the interposer INS-1. In FIG. 6,to avoid the drawing from becoming complicated, the reference charactersMBM are designated only to the leftmost microbump in the plurality ofmicrobumps. By the plurality of bumps, the interposer INS-1 is coupledto corresponding bump electrodes in the package substrate PPS-1. In FIG.6, to avoid the drawing from becoming complicated, the referencecharacters SBM are designated only to the leftmost bump in the pluralityof bumps. Similarly, by the plurality of balls, the package substratePPS-1 is coupled to corresponding ball electrodes in the board PBS. InFIG. 6, to avoid the drawing from becoming complicated, the referencecharacters SBL are designated only to the leftmost ball in the pluralityof balls.

When the print substrate PBS is seen from its first main surface PBF1side, in the first region in the print substrate PBS, as describedabove, the package substrate PPS-1, the interposer INS-1, and thesemiconductor chips (the memory semiconductor chips MCH-1 to MCH-4 andthe logic semiconductor chip LCH-1) are mounted in order. Similarly,when the print substrate PBS is seen from its first main surface PBF1side, in the second region in the print substrate PBS, the packagesubstrate PPS-2, the interposer INS-2, and the logic semiconductor chipLCH-2 are mounted in order. The first and second regions are regionsapart from each other in the print substrate PBS when viewed from thefirst main surface PFB1 side.

The package substrate PPS-2 has a configuration similar to that of thepackage substrate PPS-1. The package substrate PPS-2 is mounted so thatits second main surface PPF2 is opposed to the first main surface PBF1,and ball electrodes formed over the second main surface PPF2 of thepackage substrate PPS-2 are electrically coupled to the correspondingball electrodes PDE in the first main surface PBF1 of the printsubstrate PBS by balls. The interposer INS-2 is mounted so that itssecond main surface INF2 is opposed to the first main surface PPF1 ofthe package substrate PPS-2, and bump electrodes (not illustrated)formed over the second main surface INF2 of the interposer INS-2 areelectrically coupled to corresponding bump electrodes in the first mainsurface PPF1 of the package substrate PPS-2 by bumps. Further, the logicsemiconductor chip LCH-2 is mounted so that its second main surface CHF2is opposed to the first main surface INF1 of the interposer INS-2, andmicrobump electrodes formed over the second main surface of the logicsemiconductor chip LCH-2 are electrically coupled to correspondingmicrobump electrodes in the first main surface INF1 of the interposerINS-2 by microbumps.

Like the package substrate PPS-1, the package substrate PPS-2 has aplurality of conductive layers and a plurality of insulating layersalternately stacked. In FIG. 6, a part of a wiring pattern formed by theconductive layers in the package substrate PPS-2 is illustrated. In FIG.6, reference numerals PPL2-1 and PPL2-2 are designated to a part of thewiring pattern (signal wire). Like the interposer INS-1, the interposerINS-2 also has a plurality of conductive layers and a plurality ofinsulting layers alternately stacked. In FIG. 6, a part of a wiringpattern formed by the conductive layers in the interposer INS-2 isillustrated. In FIG. 6, reference numerals INL2-1 and INL2-2 aredesignated to a part of the wiring pattern (signal wire).

The logic semiconductor chip LCH-2 has a SerDes circuit and the SerDeshas a plurality of transmission buffer circuits and a plurality ofreception buffer circuits. The transmission buffer circuits and thereception buffer circuits are electrically coupled to microbumps formedover the second main surface CHF2 of the logic semiconductor chip LCH-2.In FIG. 6, one transmission buffer circuit in the plurality oftransmission buffer circuits of the SerDes circuit in the logicsemiconductor chip LCH-2 is illustrated with reference numerals SCB2-1and two reception buffer circuits in the plurality of reception buffercircuits are illustrated with reference numerals RCB2-1 and RCB2-2.

In the SerDes circuit in the logic semiconductor chip LCH-2, a serialsignal is supplied from the transmission buffer circuit SCB2-1 via afirst signal line to the reception buffer circuit RCB1-2 in the SerDescircuit in the logic semiconductor chip LCH-1. A serial signal issupplied from the transmission buffer circuit SCB1-1 in the SerDescircuit in the logic semiconductor chip LCH-1 to the reception buffercircuit RCB2-2 in the SerDes circuit in the logic semiconductor chipLCH-2 via a second signal line. In such a manner, high-speedtransmission/reception of the serial signal can be performed between thelogic semiconductor chips LCH-2 and LCH-1. For example, the memorysemiconductor chips MCH-1 to MCH-4 can be accessed at high speed fromthe logic semiconductor chip LCH-2.

The reception buffer circuits RCB1-1 and RCB2-1 in the logicsemiconductor chips LCH-1 and LCH-2 are, for example, coupled to anot-illustrated semiconductor chip via a not-illustrated signal line andused to receive a high-speed serial signal.

The transmission buffer circuit SCB2-1 and the reception buffer circuitRCB2-1 are electrically coupled via wiring patterns INL1-1 and INL2-1 inthe interposers INS-1 and INS-2, wiring patterns PPL1-1 and PPL2-1 inthe package substrates PPS-1 and PPS-2, and the wiring pattern PBL1 inthe print substrate PBS. Similarly, the transmission buffer circuitSCB1-2 and the reception buffer circuit RCB2-2 are electrically coupledvia wiring patterns INL1-2 and INL2-2 in the interposers INS-1 andINS-2, wiring patterns PPL1-2 and PPL2-2 in the package substrates PPS-1and PPS-2, and the wiring pattern PBL2 in the print substrate PBS.

That is, a first signal line is comprised of the wiring patterns INL1-1,INL2-1, PPL1-1, PPL2-1, and PBL1 coupled in series between the outputterminal of the transmission buffer circuit SCB2-1 and the inputterminal of the reception buffer circuit RCB1-2. A second signal line iscomprised of the wiring patterns INL1-2, INL2-2, PPL1-2, PPL2-2, andPBL2 coupled in series between the output terminal of the transmissionbuffer circuit SCB1-1 and the input terminal of the reception buffercircuit RCB2-2.

In the first embodiment, the wire width of the wiring pattern (forexample, INL1-1, INL2-1, or the like) in the interposers INS-1 and INS-2is narrower than that of the wiring pattern (for example, PPL1-1,PPL2-1, or the like) in the package substrates PPS-1 and PPS-2 and thewiring pattern (for example, PBL1 or the like) in the print substratePBS. That is, the signal density in the interposers INS-1 and INS-2 canbe made higher than that of the package substrate and the printsubstrate. Consequently, for example, like the interposer INS-1, thesemiconductor chips mounted on the same interposer can be easily coupledby a wiring pattern in the interposer.

The wire width of the wiring pattern in the print substrate PBS isthicker than that of the wiring pattern in the package substrates PPS-1and PPS-2. According to the wire width of the wiring pattern, the sizeof an electrode coupled to the wiring pattern changes. Consequently, thesize of a ball electrode formed over the first main surface PBF1 in theprint substrate PBS is larger than that of a microbump electrode coupledby a microbump. In the first embodiment, the size of a bump electrodeformed over the first main surface PPF1 of the package substrates PPS-1and PPS-2 is set to the size between the microbump electrode and theball electrode. Therefore, the electrodes can be disposed at highdensity in a state where the wire width of the wiring pattern of theprint substrate PBS, that of the package substrates PPS-1 and PPS-2, andthat of the interposers INS-1 and INS-2 are in descending order.

In this case, the size of the microbump MBM, that of the bump SBM, andthat of the ball SBL are in ascending order. The microbump MBM, the bumpSBM, and the ball SBL are deformed at the time of electrically couplingthe electrodes. Consequently, it may be understood that the comparisonof the sizes is in a state before the electrodes are coupled.

In the first embodiment, the memory semiconductor chips MCH-1 to MCH-4are disposed three-dimensionally and the logic semiconductor chip LCH-1is disposed two-dimensionally. Consequently, it can be also understoodthat the semiconductor device 6000 illustrated in FIG. 6 is a so-called2.5-D semiconductor device. The SerDes circuit described in FIG. 6 is,although not limited, a SerDes circuit having transfer speed of 56 Gbps.

The interposers INS-1 and INS-2 are, although not limited, siliconinterposers. For example, the interposers INS-1 and INS-2 may beinterposers using a glass substrate or an organic substrate.

In FIG. 6, regions DFA-1 and DFA-2 surrounded by broken lines indicateregions in which the digital filter described with reference to FIG. 1and the like is formed. In FIG. 6, MM indicates an insulator partcovering the coupling part of the semiconductor chip and the interposer.

Structure of Digital Filter

Next, the structure of the digital filter according to the firstembodiment will be described. FIG. 7 is a plan view of the broken-lineregion DFA-1 in FIG. 6 seen from the first main surface PPF1 of theinterposer PPS-1. FIG. 8 is a cross section illustrating an A-A′ sectionand a B-B′ section in FIG. 7. FIGS. 7 and 8 illustrate an example thatthe delay element DLN as a component of the digital filter 1002 iscomprised of the conductive layers formed in the interposer PPS-1.

In FIG. 7, each of INS-L10 to INS-L14 and INS-L10 to INS-L12 indicates awiring pattern (signal wire) formed by the conductive layer formed inthe interposer INS-1. Although an example will be described later withreference to FIG. 8, the interposer INS-1 has three conductive layersINS-L1 to INS-L3 isolated from one another by the insulating layers.Although not limited, the wiring patterns INS-L10 to INS-L14 are formedby the first conductive layer INS-L1 in the three conductive layers, andeach of the wiring patterns INS-L30 to INS-L32 is formed by the thirdconductive layer INS-L3.

As described with reference to FIGS. 1 to 5, the delay element DLN has asignal line to/from which a signal to be transmitted is input/output anda voltage line which extends in parallel to the signal line and to whichthe predetermined voltage Vs is supplied. In FIG. 7, the wiring patternsINS-L10 is used as the signal line to/from which the signal to betransmitted is input/output, and the wiring patterns INS-L11 and INS-L12are used as the voltage lines to which the predetermined voltage Vs issupplied. As understood from FIG. 7, each of the wiring patterns INS-L11and INS-L12 (seventh and eighth wiring patterns) has a region opposed(parallel in FIG. 7) to the wiring pattern INS-L10.

Consequently, when viewed from the first main surface INF1 of theinterposer INS-1, the wiring pattern (signal wire) INS-L10 extends inthe lateral direction in FIG. 7, and the wiring patterns (signal lines)INS-L11 and INS-L12 are parallel to the wiring pattern INS-L10 in planview and extend in the lateral direction as illustrated in FIG. 7. Theother end of each of the wiring patterns INS-L10, INS-L11, and INS-L12is coupled to the wiring pattern INS-L13 extending in the verticaldirection in FIG. 7 and disposed so as to be perpendicular to each ofthe wiring patterns INS-L10, INS-L11, and INS-L12. One end of each ofthe wiring patterns INS-L11 and INS-L12 is coupled to the wiring patternINS-L14 extending in the vertical direction in FIG. 7 and disposed so asto be perpendicular to each of the wiring patterns INS-L10 and INS-L12.

The wiring pattern INS-L14 is coupled to wiring patterns INS-L31 andINS-L32 formed by the third conductive layer INS-L3 via contacts CT2buried with a conductive material so as to couple the conductive layers.The wiring pattern INS-L14 is coupled to microbump electrodes(hereinbelow, electrodes will be also called pads) INS-MPD1 and INS-MPD2formed over the first main surface INF1 of the interposer INS-1 via thecontacts CT2. On the other hand, one end of the wiring pattern INS-L10is coupled to the wiring pattern INS-L30 formed by the third conductivelayer INS-L3 via the contacts CT2. One end of the wiring pattern INS-L10is coupled to a microbump electrode INS-MPDS formed over the first mainsurface INF1 of the interposer INS-1 via the contacts CT2.

As will be described later, to the wiring patterns INS-L31 and INS-L32,the predetermined voltage Vs (for example, the ground voltage of thecircuit) is supplied via a wiring pattern formed in the packagesubstrate PPS-1 (FIG. 6) and the print substrate PBS (FIG. 6). To thewiring pattern INS-L30, a transmission signal is supplied from the logicsemiconductor chip LCH-2 via a signal wire (wiring pattern) formed inthe package substrate PPS-1 and the print substrate PBS.

The microbump electrodes INS-MPD1 and INS-MPD2 are coupled tocorresponding microbump electrodes in the second main surface CHF2 ofthe logic semiconductor chip LCH-1 by the microbumps MPM (in FIG. 7,MBM-G1 and MBM-G2). With the configuration, the predetermined voltage Vsis supplied to the logic semiconductor chip LCH-1. The predeterminedvoltage Vs is used as a voltage for operating circuit blocks formed inthe logic semiconductor chip LCH-1 (for example, the transmission buffercircuit SCB1-1 and the reception buffer circuit RCB1-2 illustrated inFIG. 6).

The microbump electrode INS-MPDS is coupled to a corresponding microbumpelectrode over the second main surface CHF2 of the logic semiconductorchip LCH-1 by the microbump MBM (in FIG. 7, MBM-S1). In this case, thecorresponding microbump electrode is coupled to the input terminal ofthe reception buffer circuit RCB1-2. With the configuration, atransmission signal from the logic semiconductor chip LCH-2 is suppliedto the reception buffer circuit RCB1-2 provided in the logicsemiconductor chip LCH-1.

In the example illustrated in FIG. 7, in the plan view, thepredetermined voltage Vs is supplied to the wiring pattern INS-L10to/from which a signal to be transmitted is input/output, and the wiringpattern INS-L10 is surrounded by the wiring patterns INS-L11 to LNS-L14formed by the conductive layer which is the same as that of the wiringpattern INS-L10. It can also prevent leakage of electromagnetic fieldgenerated when the signal input to the wiring pattern INS-L10 changes.

Next, with reference to FIG. 8, the structure of the digital filterDFA-1 will be described. FIG. 8 illustrates the section of not only theinterposer INS-1 as a component of the digital filter DFA-1 but also apart of the logic semiconductor chip LCH, a part of the packagesubstrate PPS-1, and a part of the print substrate PBS. The partillustrated in FIG. 8 is only the part related to the digital filterDFA-1 and the other part is not illustrated. FIG. 8 is the A-A′ sectionviewed from the B-B′ side in FIG. 7. Consequently, in FIG. 8, the A-A′section and the B-B′ section are partially overlapped.

The print substrate PBS has a plurality of conductive layers and aplurality of insulating layers alternately stacked up. In FIG. 8, tofacilitate the explanation, the print substrate PBS having only aninsulating layer PBO and one conductive layer PBL stacked on theinsulting layer PBO is illustrated. By the conductive layer PBLillustrated in FIG. 8, the wiring pattern PBL1 illustrated in FIG. 6 isformed. In FIG. 8, the ball electrodes SBL described with reference toFIG. 6 are not illustrated.

The package substrate PPS-1 has, although not limited, four conductivelayers PPS-L1 to PPS-L4 which are isolated from one another byinsulating layers.

FIG. 8 illustrates, for explanation, a part of the package substratePPS-1 related to the bump MBM-S1 described in FIG. 7. Over the secondmain surface PPF2 of the package substrate PPS-1, as described withreference to FIG. 6, the ball electrode is formed. In FIG. 8, the ballelectrode is indicated as reference characters PPS-LPD. The ballelectrode PPS-LPD is coupled to the wiring pattern PBL via thenot-illustrated ball electrode SBL by the ball SBL. On the first mainsurface PPF1 of the package substrate PPS-1, as described in FIG. 6, thebump electrode is formed. In FIG. 8, the bump electrode is indicated byreference characters PPS-SPD.

In FIG. 8, PPS-L4(R), PPS-L3(R), PPS-L2(R), and PPS-L1(R) indicatepredetermined wiring patterns formed by the conductive layers PPS-L1 toPPS-L4. The reference character (R) indicates a wiring pattern relatedto the wiring pattern INS-L10 illustrated in FIG. 7, and the referencecharacters before the reference character (R) indicate the conductivelayer forming the wiring pattern. For example, PPS-L4(R) indicates awiring pattern related to the wiring pattern INS-L10 formed by thefourth conductive layer. The wiring patterns PPS-L3(R) to PPS-L1(R) areexpressed by the same notation system.

Openings are formed in the insulating layers interposed between thewiring patterns PPS-L1(R) to PPS-L4(R) and filled with a conductivematerial, thereby forming contacts CT3. Via the contacts CT3, asillustrated in FIG. 8, the wiring patterns PPS-L1(4) to PPS-L4(4) arecoupled to one another. Via the contact CT3, the wiring patternPPS-L1(R) and the ball electrode PPS-LPD are coupled. Similarly, via thecontact CT3, the wiring pattern PPS-L4(R) and the bump electrode PPS-MPDare coupled. As a result, by the wiring patterns PPS-L1(R) to PPS-L4(R)and the contacts CT3, the ball electrode PPS-LPD and the bump electrodePPS-MPD are electrically coupled and the wiring pattern (signal line)PPL1-1 illustrated in FIG. 6 is formed.

The interposer INS-1 is, although not limited, a silicon interposer.That is, the interposer INS-1 is comprised of a silicon substrate SSBIand a wiring layer HSB-I having a plurality of conductive layers formedon the silicon substrate SSBI by the known semiconductor manufacturingtechnology. In the first embodiment, the wiring layer HSB-I has thethree conductive layers INS-L1 to INS-L3. Obviously, between theadjacent conductive layers, an insulating layer is interposed. Asdescribed in FIG. 6, the bump electrode is formed over the second mainsurface INF2 of the interposer INS-1, and the microbump electrode isformed over the first main surface INF1. In FIG. 8, the bump electrodeformed over the second main surface INF2 is indicated by referencecharacters INS-SPD, and the microbump electrodes formed over the firstmain surface INF1 are indicated by reference numerals INS-MPD2 andINS-MPDS (refer to FIG. 7).

In FIG. 8, INS-L1(R) to INS-L3(R) indicate wiring patterns formed by theconductive layers INS-L1 to INS-L3, and INS-L1(V) to INS-L3(V) alsoindicate wiring patterns formed by the conductive layers INS-L1 toINS-L3. The wiring patterns are also expressed by the notation systemused in the description of the package substrate. Specifically, thereference character (R) indicates a wiring pattern related to the wiringpattern INS-L10 illustrated in FIG. 7, and the reference charactersbefore the reference character (R) indicate the conductive layer formingthe wiring pattern. The reference character (V) indicates a wiringpattern related to the wiring pattern INS-L12 illustrated in FIG. 7, andthe characters before the reference character (V) indicate theconductive layer forming the wiring pattern.

In FIG. 8, the A-A′ section and the B-B′ section illustrated in FIG. 7are drawn overlapped. First, the A-A′ section will be described. In theleft part in FIG. 8, by a contact CT2S formed by filling an openingformed in the silicon substrate SSB-I with a conductive material, thebump electrode TNS-SPD (second electrode) is coupled to the wiringpattern INS-L1(R). The wiring pattern INS-L1(R) is coupled to the wiringpattern INS-L2(R) in the second layer via the contact CT2, and thewiring pattern INS-L2(R) in the second layer is coupled to the wiringpattern INS-L3(R) in the third layer via the contact CT2. The wiringpattern INS-L3(R) in the third layer corresponds to the wiring patternINS-L30 illustrated in FIG. 7. The wiring pattern INS-L3(R) in the thirdlayer corresponds to the wiring pattern INS-L30 illustrated in FIG. 7.That is, in plan view of FIG. 7, a part of the wiring pattern INS-L3(R)in the third layer has a plane shape drawn as the wiring patternINS-L30.

The wiring pattern INS-L3(R) is coupled to the microbump electrodeINS-MPDS (first electrode) via the contact CT2. In the right part inFIG. 8, the wiring pattern INS-L3(R) is coupled to the wiring patternINS-L2(R) which is hatched via the contact CT2, and the wiring patternINS-L2(R) in the second layer is coupled to the wiring pattern INS-L1(R)(fourth wiring pattern) which is hatched via the contact CT2. Thehatched wiring pattern INS-L1(R) in the first layer corresponds to thewiring pattern INS-L10 illustrated in FIG. 7. That is, in plan view, thehatched wiring pattern INS-L1(R) has a plane shape as illustrated by thewiring pattern INS-L10 in FIG. 7.

Next, the B-B′ section illustrated in FIG. 7 will be described. Althoughnot illustrated in FIG. 8, over the second main surface INF2 of theinterposer INS-1, a bump electrode to which the predetermined voltage Vsis supplied is formed. The wiring pattern INS-L1(V) in the first layerillustrated in the left part of FIG. 8 is coupled to a not-illustratedbump electrode via the contact CT2S. The wiring pattern INS-L1(V) iscoupled to the wiring pattern INS-L2(V) via the contact CT2, and thewiring pattern INS-L2(V) is coupled to the wiring pattern INS-L3(V) viathe contact CT2. The wiring pattern INS-L3(V) in the third layercorresponds to the wiring pattern INS-L32 illustrated in FIG. 7. Thatis, the plane shape of a part of the wiring pattern INS-L3(V) in thethird layer is the shape of the wiring pattern INS-L32 illustrated inFIG. 7.

The wiring pattern INS-L3(V) is coupled to the hatched wiring patternINS-L2(V) via the contact CT2 in the right part of FIG. 8 and, further,the wiring pattern INS-L2(V) is coupled to the wiring pattern INS-L1(V)(sixth wiring pattern) which is hatched via the contact CT2. The hatchedwiring pattern INS-L1(V) in the first layer corresponds to the wiringpattern INS-L12 illustrated in FIG. 7. That is, the plane shape of thehatched wiring pattern INS-L1(V) is the shape of the wiring patternINS-L12 illustrated in FIG. 7.

The wiring pattern INS-L3(V) in the third layer is coupled to themicrobump electrode INS-MPD2 via the contact CT.

Like the bump electrode INS-SPD illustrated in FIG. 8, the bumpelectrode which is not illustrated in FIG. 8 is coupled to the wiringpattern of the print substrate SBP via the package substrate PPS-1, andthe predetermined voltage Vs is supplied via the wiring pattern of theprint substrate SBP. Like the contact CT3, the contact CT2 is formed byproviding an opening in an insulating layer interposed between theconductive layers and filling the opening with a conductive material.

In FIG. 8, it can be regarded that the wiring pattern INL1-1 illustratedin FIG. 6 is comprised of the wiring patterns INS-L1(R), INS-L2(R), andINS-L3(R), the contact CT2 coupling the wiring patterns, and the contactCT2S illustrated on the left side.

The logic semiconductor chip LCH-1 has a semiconductor substrate, forexample, the silicon substrate SSB in which a semiconductor regionconfiguring an element and the like is formed and the wiring layer HSBformed on the main surface of the silicon substrate SSB. The wiringlayer HSB has a plurality of conductive layers and a plurality ofinsulating layers which are stacked alternatively. Although not limited,in the first embodiment, the wiring layer HSB has three wiring layersLCH-L1 to LCH-L3. In FIG. 8, SS indicates a semiconductor region formedin the semiconductor substrate SSB. The semiconductor region SScorresponds to the input terminal of the reception buffer circuit RCB1-2(FIG. 6).

In FIG. 8, LCH1-L1(R), LCH-L2(R), LCH-L3(R) and LCH-L3 indicate wiringpatterns. The wiring patterns LCH1-L1(R), LCH-L2(R), and LCH-L3(R) aredrawn by the same notation system as that of the wiring patterns formedin the interposer INS-1. That is, the reference character (R) indicatesa wiring pattern related to the wiring pattern INS-L10 illustrated inFIG. 7, and the reference characters before the reference character (R)indicate the conductive layer in which the wiring pattern is formed.

As described with reference to FIG. 6, a plurality of microbumpelectrodes are formed over the second main surface CHF2 of the logicsemiconductor chip LCH-1. In FIG. 8, LCH-PD2 and LCH-PD3 indicatemicrobump electrodes formed on the second main surface CHF2.

The microbump electrode INS-MPDS on the interposer INS-1 is coupled tothe microbump electrode LCH-PD3 formed on the second main surface CHF2by the microbump MBM-S1. The microbump electrode LCH-PD3 is coupled tothe wiring pattern LCH-L3(R) (fifth wiring pattern) in the third layervia the contact CT1. The wiring pattern LCH-L3(R) is coupled to thewiring pattern LCH-L2(R) in the second layer via the contact CT1, andthe wiring pattern LCH-L2 is coupled to the wiring pattern LCH-L1(R) inthe first layer. Further, the wiring pattern LCH-L1(R) is coupled to thesemiconductor region SS via the contact CT1. The wiring patternINS-L3(R) illustrated in FIG. 8 corresponds to the wiring patternINS-L30 illustrated in FIG. 7, and the microbump electrode INS-MPDS andthe bump electrode INS-SPD are coupled via the wiring pattern INS-L3(R).

The microbump electrode LCH-PD2 is coupled to the microbump electrodeINS-MPD2 by the microbump MBM-G2, and the microbump electrode INS-MPD2is coupled to a wiring pattern LCH-L3(O) via the contact CT1. The wiringpattern LCH-L3(O) is used, for example, as a wire supplying thepredetermined voltage Vs to the reception buffer circuit RCB1-2. Thecontact CT1 is formed by, like the contact CT2, forming an opening inthe insulating layer between the conductive layers and filling theopening with a conductive material.

With the configuration, a transmission signal from the logicsemiconductor chip LCH-2 (FIG. 6) is transmitted to the input terminalof the reception buffer circuit RCB1-2 via the wiring pattern in theprint substrate PBS, the wiring pattern in the package substrate PPS-1,and the wiring pattern in the interposer INS-1.

A wiring pattern L10 illustrated in FIG. 7 (first wiring pattern: inFIG. 8, the hatched wiring pattern INS-L1(R)) corresponds to the delayelement DLN described with reference to FIG. 1 and the like, and thewiring patterns INS-L11 and INS-L12 illustrated in FIG. 7 (second wiringpatterns: in FIG. 8, the hatched wiring pattern INS-L1(V)) extend inparallel to the delay element DLN and become wiring patterns (voltagewires) to which the predetermined voltage Vs is supplied. When a signalis input, the wiring pattern INS-L10 is set so that a round-trip delayas delay time of a signal which is output becomes a fraction of aninteger of the data width interval UT (round-trip signal delay UT/m).

In FIG. 6, a transmission signal output from the transmission buffercircuit SCB2-1 in the logic semiconductor chip LCH-2 is transmitted tothe wiring pattern PBL1 in the print substrate PBS via the wiringpattern INL2-1 in the interposer INS and the wiring pattern PPL2-1 inthe package substrate PPS-2. A transmission signal transmitted to thewiring pattern PBL1 in the print substrate PBS propagates through thewiring pattern PBL1, is transmitted to the wiring pattern PPL1-1 in thepackage substrate PPS-1 and the wiring pattern INL1-2 in the interposerINS-1, and is transmitted to the microbump electrode LCH-PD3 illustratedin FIG. 8.

The transmission signal transmitted to the microbump electrode LCH-PD3is transmitted to the input terminal (for example, the semiconductorregion SS) of the reception buffer circuit RCB1-2, amplified, andprocessed. The transmission signal transmitted to the microbumpelectrode LCH-PD3 is input also to one end of the wiring pattern INS-L10as a component of the delay element DLN. The other end of the wiringpattern INS-L10 is coupled to the predetermined voltage Vs.Consequently, as described with reference to FIG. 1 and the like, areflecting wave is generated and output from one end of the wiringpattern to the microbump electrode LCH-PD3.

The wiring patterns INL2-1, PPL2-1, PBL1, PPL1-1, and INL1-2 are coupledin series to configure the signal line 1000 described with reference toFIG. 1 and the like. The transmission signal deteriorated by a loss inthe signal line 1000 is restored (shaped) by the output signal(reflecting wave) output from one end of the wiring pattern INS-L10, andthe resultant signal is supplied to the input terminal of the receptionbuffer circuit RCB1-2.

In the first embodiment, one end of the wiring pattern INS-L10 as acomponent of the delay element DLN functions as an input/output terminaland the other end is coupled to the predetermined voltage Vs (forexample, the ground voltage of the circuit). The wiring patterns(voltage lines) INS-L11 and INS-L12 as components of the delay elementDLN, to which the predetermined voltage Vs is supplied, are disposed soas to extend in parallel to the wiring pattern (signal line) INS-L10while sandwiching the wiring pattern INS-L10 therebetween in plan view.That is, the delay element DLN has a structure of a so-called co-planarwaveguide. From the viewpoint that the predetermined voltage Vs issupplied to the other end of each of the wiring patterns INS-L10 toINS-L12, it can be regarded that the signal line (wiring patternINS-L10) as a component of the delay element DLN and the other end ofeach of the voltage lines (wiring patterns INS-L11 and INS-L12) areshort-circuited.

In the first embodiment, the thicknesses of the conductive layers in theinterposer INS-1 are the same. As there is a condition that the signalloss (electric resistance ratio) per unit length of the delay elementDLN is smaller than that of the signal line, it is sufficient to makeline width BLD1 (FIG. 7) of the signal line (wiring pattern INS-L10) asa component of the delay element DLN and line width BLD2 (FIG. 7) of thevoltage line (wiring patterns INS-L11 and INS-L12) thinner than linewidth BLS (FIG. 7) of the signal line (for example, wiring patternINS-L30).

FIGS. 7 and 8 have been described using the wiring patterns INL1-1 andPPL1-1 illustrated in FIG. 6 as an example. The wiring patterns INL1-2,PPL1-2, INL2-1, PPL2-1, INL2-2, and PPL2-2 illustrated in FIG. 6 aresimilar. In this case, a wiring pattern similar to the wiring pattern asa component of the delay element DLN described with reference to FIGS. 7and 8 may be coupled to the input terminal of each of the receptionbuffer circuits or coupled to the output terminal of each of thetransmission buffer circuits. The wiring pattern as a component of eachof the delay elements DLN may be coupled to both of the input terminalof the reception buffer circuit and the output terminal of thetransmission buffer circuit.

As illustrated in FIGS. 7 and 8, in the case of forming the wiringpattern INS-L30 (third wiring pattern: INS-L3(R)) as a component of thesignal line and the wiring patterns INS-L10 (INS-L1(R)), IND-L11, andINS-L12 (INS-L1(V)) configuring the delay element DLN by differentconductive layers, the extension direction of the wiring pattern as acomponent of the signal line and that of the wiring patterns configuringthe delay element DLN can be determined arbitrarily as long as they arenot in contact.

Details of Delay Element

Although limited, the wiring pattern INS-L10 as a component of the delayelement DLN is formed by a thin metal wiring layer, and an equivalentcircuit of the delay element DLN is expressed by FIG. 1B. By the skineffect of the wiring pattern INS-L10, resistance R in the equivalentcircuit illustrated in FIG. 1B becomes large. As a result, the delayelement becomes to have large signal attenuation. In place of increasingthe resistance R, the conductance G illustrated in FIG. 1B may beincreased. In this case, a signal flowing in the predetermined voltageVs becomes large, and the signal attenuation becomes large.

As described with reference to FIG. 1 and the like, a transmissionsignal from the signal line 1000 is distributed to the digital filter1002 and the reception buffer circuit 1003 at the part (node WRN) of thewired-OR coupling. The equation (7) illustrated in FIG. 3 expresses thesignal distribution. In the equation (7), Z₀ expresses impedance of thepart of the wired-OR coupling (for example, the node WRN in FIG. 1A)when the delay element DLN is not coupled. In the equation (7), Z_(c)indicates the impedance of the delay element DLN.

As illustrated in the equation (7), the impedance Z_(c) changesaccording to the inductance L and capacitance C illustrated in FIG. 1B.The inductance L and the capacitance C can be changed by changing thedistance (interval) BLL between the wiring pattern INS-L10 illustratedin FIG. 7 and each of the wiring patterns INS-L11 and INS-L12. That is,when the interval BLL illustrated in FIG. 7 is enlarged, the inductanceL increases, and the capacitance C decreases. On the contrary, when theinterval BLL is narrowed, the inductance L decreases, and thecapacitance C increases. By setting the interval BLL to a desired value,a state where impedance Z_(c)<impedance Z₀ and a state where impedanceZ_(c)>impedance Z₀ can be formed. By setting the states of theimpedances Z_(c) and Z₀, the value of a distribution ratio of a signal(signal distribution ratio) can be controlled by the equation (7).

That is, at the time of performing restoration (shaping), the value ofthe signal output from the delay element DLN to the part of the wired-ORcoupling can be controlled by the interval BLL.

In FIG. 7, the delay element DLN is comprised of the signal line (wiringpattern INS-L10) and the voltage lines (wiring patterns INS-L11 andINS-L12) disposed so as to sandwich the signal line, and thepredetermined voltage Vs is supplied to the voltage lines. However, thevoltage lines may be disposed on only one of the sides. In this case, onthe side that the voltage line is not provided, an electromagnetic fieldmay be leaked. Consequently, the configuration is not suitable tohigh-speed signal transmission. However, in the case of disposing thevoltage line only one of the sides to reduce the area or the like, it isdesirable to set the state of impedance Z_(c)<impedance Z₀ by narrowingthe interval BLL.

Eye Pattern

FIGS. 9A and 9B are diagrams illustrating eye patterns in thesemiconductor device according to the first embodiment. In FIGS. 9A and9B, the horizontal axis indicates time, and the vertical axis indicatessignal voltage. FIGS. 9A and 9B illustrate the case where an FR4 (FlameRetardant Type 4) substrate is used as a print substrate, the logicsemiconductor chips LCH-1 and LCH-2 as illustrated in FIG. 6 are mountedover the print substrate so that the interval of the logic semiconductorchips LCH-1 and LCH-2 becomes four inches, and a signal in the NRZ formis supplied from the logic semiconductor chip LCH-2 to the logicsemiconductor chip LCH-1 via the wiring pattern of the print substrateat transfer speed of 56 Gbps. As the interposer, a silicon interposer isused. FIG. 9A is a diagram of an eye pattern drawn by overlapping signalvoltage waveforms at the input terminal of the reception buffer circuitwhen the digital filter 1002 is coupled to the reception buffer circuitof the logic semiconductor chip LCH-1. On the other hand, FIG. 9B is adiagram of an eye pattern drawn by overlapping voltage waveforms at theinput terminal of the reception buffer circuit in a state where thedigital filter 1002 is not coupled to the reception buffer circuit ofthe logic semiconductor chip LCH-1.

In the case of comparing FIGS. 9A and 9B, in FIG. 9B, the voltagewaveform at the input terminal fluctuates so that it becomes difficultto specify the eye pattern. In contrast, in FIG. 9A, the eye pattern canbe recognized, so that a transmission signal can be specified.

FIGS. 10A and 10B are diagrams, like FIGS. 9A and 9B, illustrating eyepatterns in the case where the digital filter is coupled to the inputterminal of the reception buffer circuit and the case where the digitalfilter is not provided. The points different from FIGS. 9A and 9B arethat a glass interposer is used as the interposer, the interval betweenthe logic semiconductor chips is 6 inches, and the transmission speed is31.25 Gbps. FIG. 10A illustrates the eye pattern when the digital filter1002 is coupled. FIG. 10B illustrates the eye pattern when the digitalfilter 1002 is not provided. In the case of comparing FIGS. 10A and 10B,like FIGS. 9A and 9B, when the digital filter 1002 is not provided, itis difficult to specify the eye pattern. On the contrary, in FIG. 10A,the eye pattern can be specified, and a transmission signal can bespecified.

That is, by providing the digital filter 1002 comprised of a passiveelement not an active element such as a transistor, while suppressingincrease in power consumption, a signal (data) can be specified.

Modification

In FIG. 8, the wiring pattern INS-L30 (FIG. 7) forming the signal lineand the wiring pattern INS-L10 (FIG. 7) as a component of the delayelement DLN are formed by different conductive layers. That is, thewiring pattern INS-L30 is, as illustrated in FIG. 8, the wiring patternINS-L3(R) in the third layer, and the wiring pattern L10 is the wiringpattern INS-L1(R) in the first layer which is hatched. In FIG. 8, it canbe also regarded that the wiring pattern INS-L2(R) in the second laterwhich is hatched is a part of the wiring pattern forming the delayelement DLN.

On the other hand, in the modification, the wiring pattern forming thesignal line and the wiring pattern forming the delay element DLN areformed in the same layer. FIG. 11 is a plan view of a semiconductordevice according to the modification. FIG. 12 is a cross sectionillustrating an A1-A1′ section and a B1-B1′ section in FIG. 11.

FIG. 11 is similar to FIG. 7, and FIG. 12 is similar to FIG. 8. Thepoint that FIGS. 11 and 12 are different from FIGS. 7 and 8 is that thewiring pattern forming the signal line and the wiring pattern formingthe delay element DLN are formed by the same layer. Only the differentpoint will be mainly described here.

In FIG. 11, the wiring pattern INS-L10 as a component of the signal lineand the wiring pattern INS-L10 as a component of the delay element DLNare integrally formed by the same conductive layer. In the modification,the wiring patterns INS-L30 and INS-L10 are formed by the thirdconductive layer INS-L3. In FIG. 11, in plan view, the wiring patterndisposed on the left side of the region overlapped with the microbumpelectrode INS-MPDS is the wiring pattern INS-L30 as a component of thesignal line, and the wiring pattern disposed on the right side of themicrobump electrode INS-MPDS is the wiring pattern INS-L10 as acomponent of the delay element DLN.

In the example of FIG. 11, the line width BLS of the wiring patternINS-L30 and the line width BLD1 of the wiring pattern INS-L10 aredifferent. The line width BLD1 is narrower than the line width BLS.Consequently, in the wiring patterns integrally formed, using the regionat which the line width changes as a border, the wider wiring patternmay be distinguished as the wiring pattern INS-L30 and the narrowerwiring pattern may be distinguished as the wiring pattern INS-L10.

In FIG. 11, the wiring patterns INS-L11 and INS-L12 function as thevoltage lines of the delay element DLN. The wiring patterns INS-L11 andINS-L12 are also formed by the same conductive layer as that of thewiring patterns INS-L31 and INS-L32 for supplying the predeterminedvoltage Vs. In the modification, the wiring patterns INS-L11 and INS-L12are formed by the third conductive layer INS-L3 which is the same asthat of the wiring patterns INS-L10 and INS-L30. Specifically, thewiring pattern INS-L11 is formed integrally with the wiring patternINS-L31, and the wiring pattern INS-L12 is formed integrally with thewiring pattern INS-L32.

In the example of FIG. 11, the line width of the wiring pattern INS-L31and that of the wiring pattern INS-L11 are different. Similarly, theline width of the wiring pattern INS-L32 and that of the wiring patternINS-L12 are different. Specifically, the line width BLD2 of the wiringpatterns INS-L11 and INS-L12 is narrower than the line width of thewiring patterns INS-L31 and INS-L32. Consequently, using the region atwhich the line width changes as a border, the wider regions can bedistinguished as the wiring patterns INS-L31 and INS-L32 and thenarrower regions can be distinguished as the wiring pattern INS-L11 andINS-L12.

The other ends of the wiring patterns INS-L10 to INS-L12 are coupled tothe wiring pattern INS-L13. The wiring pattern INS-L13 is also formed bythe third conductive layer INS-L3 which is the same as that of thewiring patterns INS-L10 to INS-L12. Consequently, it can be regardedthat the wiring patterns INS-L10 to INS-L13 and INS-L30 to INS-L32 areformed integrally. In FIG. 7, one end of the wiring pattern INS-L11 andthat of the wiring pattern INS-L12 are coupled to each other by thewiring pattern INS-L14. In the modification illustrated in FIG. 11, thewiring pattern INS-L14 is not provided, and one end of the wiringpattern INS-L11 and that of the wiring pattern INS-L12 are separated.

In FIG. 12, INS-L3(R) indicates the wiring patterns INS-L30 and INS-L10in the A1-A1′ section, and INS-L3(V) indicates the wiring patternsINS-L32 and INS-L12 in the B1-B1′ section. As illustrated in FIG. 12,each of the wiring patterns INS-L10, INS-L30, INS-L12, and INS-L32 isformed by the third conductive layer.

As described above, in the case of forming the wiring pattern as acomponent of the signal line and the wiring pattern as a component ofthe delay element DLN by the same conductive layer, it is sufficient tomake the wiring pattern as a component of the signal line extend fromthe microbump electrode INS-MPDS and to change the line width in theextended region.

In the modification, even the number of conductive layers forming theinterposer INS-1 is small, the digital filter 1002 can be configured.

In the first embodiment, the delay element DLN has the signal lineto/from which the signal is input/output and the voltage line whichextends in parallel to the signal line and to which the predeterminedvoltage Vs is supplied. In other words, it can be regarded that thedelay element DLN is comprised of delay lines. In this case, a signalloss amount per unit length of the signal line and the voltage line isset to be larger than that of the signal line. In the first embodiment,the boundary length in section of each of the signal wire and thevoltage wire is set to be smaller than that of the signal line. Todecrease the boundary length in section, the thickness of the signalwire, that of the voltage wire, and that of the signal line are set tothe same, and the line width of the signal wire and that of the voltagewire is set to be narrower than that of the signal line.

Second Embodiment

FIGS. 13 and 14 are diagrams illustrating the structure of asemiconductor device according to a second embodiment. FIG. 13 is a planview illustrating a plane of the semiconductor device. FIG. 14 is across section illustrating an A2-A2′ section and a B2-B2′ section inFIG. 13.

In the first embodiment, in plan view from the first main surface INF1of the interposer INS-1, the signal wire (for example, the wiringpattern INS-L10 in FIG. 7) and the voltage wire (for example, the wiringpattern INS-L12 in FIG. 7) configuring the delay element DLN aredisposed so as to extend in parallel. On the other hand, in the secondembodiment, the signal wire and the voltage wire configuring the delayelement DLN are disposed so as to be overlapped in plan view. That is,the signal wire and the voltage wire are disposed so as to be stacked inthe vertical direction in the interposer INS-1.

FIG. 13 is similar to FIG. 7, and FIG. 14 is similar to FIG. 8. Only thedifferent points will be described here.

As illustrated in FIG. 13, in the region of the A2-A2′ section, thewiring pattern INS-L10 and the wiring pattern INS-L12 (ninth wiringpattern) configuring the delay element DLN are overlapped. As will bedescribed later with reference to FIG. 14, the wiring pattern INS-L10 asa component of the signal line to/from which a signal is input/output isformed by the first conductive layer INS-L1, and the wiring patternINS-L12 to which the predetermined voltage Vs is supplied is formed bythe third wiring layer INS-L3. One end of the wiring pattern INS-L10 iscoupled to the microbump electrode INS-MPDS and the wiring patternINS-L30 as a component of the signal line, and the other end of thewiring pattern INS-L10 is coupled to the wiring pattern INS-L12 via thewiring pattern INS-L16 in the second layer.

The other end of the wiring pattern INS=L12 is coupled to the wiringpatterns INS-L14 and INS-L15 in the third layer. The wiring patternINS-L14 is coupled to the microbump electrode INS-MPD1 and the wiringpattern INS-L31, and the wiring pattern INS-L15 is coupled to themicrobump electrode INS-MPD2 and the wiring pattern INS-L32. Since thewiring patterns INS-L12, INS-L14, and INS-L15 are formed by the wiringlayer INS-L3 in the third layer, the wiring patterns can be integrallyformed.

As illustrated in FIG. 13, the line width BLD1 of the wiring patternINS-L10 to/from which a signal is input/output is narrower than the linewidth BLD2 of the wiring pattern INS-L12 to which the predeterminedvoltage Vs is supplied.

Like in FIG. 7, the predetermined voltage Vs is supplied to the wiringpatterns INS-L31 and INS-L32, and a transmission signal is supplied tothe wiring pattern INS-L30. The microbump electrodes INS-MPD1, INS-MPD2,and INS-MPDS are coupled to the microbump electrode of the logicsemiconductor chip by the microbumps MBM-G1, MBM-S1, and MBM-G2.

In FIG. 14, the A2-A2′ section and the B2-B2′ section in FIG. 13 areoverlapped. First, the structure related to the A2-A2′ section will bedescribed. In FIG. 14, the print substrate PBS, the package substratePPS-1, the ball SBL, and the bump SMB are the same as those in FIG. 8.In FIG. 14, the wiring patterns INS-L1(R) and INS-L2(R) illustrated inthe left part of the interposer INS-1 are also the same as those in FIG.8.

When FIGS. 13 and 7 are compared, in the second embodiment, themicrobump electrode INS-MPDS is disposed on the left side of themicrobump electrodes INS-MPD1 and INS-MPD2. Accordingly, in FIG. 14, themicrobump electrode INS-MPDS is disposed on the left side of themicrobump electrode INS-MPD2. The wiring pattern INS-L2(R) is coupled tothe wiring pattern INS-L3(R) in the third layer via the contact CT2, andthe wiring pattern INS-L3(R) is coupled to the microbump electrodeINS-MPDS via the contact CT2 and also coupled to the hatched wiringpattern INS-L2(R) via the contact CT2. The hatched wiring patternINS-L2(R) is coupled to one end of the hatched wiring pattern INS-L1(R)in the first layer via the contact CT2.

The hatched wiring pattern INS-L1(R) extends in the lateral direction inFIG. 14, and the other end of the hatched wiring pattern INS-L1(R) iscoupled to the wiring pattern INS-L2(V) in the second layer via thecontact CT2. The hatched wiring pattern INS-L1(R) corresponds to thewiring pattern INS-L10 illustrated in FIG. 13. The wiring patternINS-L2(V) corresponds to the wiring pattern INS-L16 illustrated in FIG.13.

The wiring pattern INS-L2(V) is coupled to the other end of the wiringpattern INS-L3(V) in the first layer via the contact CT2, and one end ofthe wiring pattern INS-L3(V) is coupled to the microbump electrodeINS-MPD2 via the contact CT2. The wiring pattern INS-L3(V) in the firstlayer corresponds to the wiring pattern INS-L12 illustrated in FIG. 13.That is, the wiring pattern INS-L3(V) extends in parallel to the wiringpattern INS-L1(R) in the first layer and, in plan view, is disposed soas to cover the wiring pattern INS-L1(R).

The microbump electrode INS-MPDS is coupled to the microbump electrodeLCH-PD3 by the microbump MBM-S1. Like FIG. 8, the microbump electrodeLCH-PD3 is coupled to the semiconductor region SS in the receptionbuffer circuit via the wiring patterns LCH-L3(R) to LCH-L1(R) and thecontact CT1.

The microbump electrode INS-MPD2 is coupled to the microbump electrodeLCH-PD2 by the microbump MBM-G2 in the B2-B2′ section in FIG. 13, andthe microbump electrode LCH-PD2 is coupled to the wiring patternLCH-L3(O) in the logic semiconductor chip LCH-1. In FIG. 14, the wiringpattern INS-L32 illustrated in FIG. 13 is omitted.

Also in the second embodiment, a signal from the wiring pattern INS-L30as a component of the signal line 1000 is input/output to/from one endof the wiring pattern (signal wire) INS-L10 as a component of the delayelement DLN. The other end of the wiring pattern INS-L10 is coupled tothe predetermined voltage Vs. With the configuration, the waveform canbe restored (shaped) with low consumption power. As illustrated in FIG.13, the line width BLD1 of the signal line (wiring pattern INS-L10) as acomponent of the delay element DLN and the voltage wire (wiring patternINS-L12) is narrower than the line width BLS of the wiring patternINS-L30 as a component of the signal line.

An equivalent circuit of the delay element DLN (microstrip delayelement) illustrated in FIGS. 13 and 14 has the configurationillustrated in FIG. 1B. When the line width BLD1 of the wiring patternINS-L10 and the line width BLD2 of the wiring pattern INS-L12illustrated in FIG. 13 are increased, in the equivalent circuitillustrated in FIG. 1B, the inductance L decreases, the capacitance Cincreases, and the resistance R decreases. On the contrary, when theline widths BLD1 and BLD2 are narrowed, the inductance L increases, thecapacitance C decreases, and the resistance R increases. In FIG. 14, bychanging the interval BRV between the wiring pattern INS-L3(V) (INS-L10)and the hatched wiring pattern INS-L1(R) (INS-L12) which extend inparallel to each other, the inductance L and the capacitance Cillustrated in FIG. 1B can be increased/decreased. For example, byincreasing the interval BRV, the inductance L can be increased, and thecapacitance C can be decreased. As a result, in a manner similar to thefirst embodiment, the signal loss amount can be set to a desired value.

In the second embodiment, the signal wire and the voltage wireconfiguring the delay element DLN are disposed stereoscopically. Sincethe line width of the signal wire and the voltage wire is narrower thanthat of the wiring pattern as a component of the signal line 1000, thearea occupied by the delay element DLN can be reduced.

Although FIG. 14 illustrates that the delay element DLN is configured byusing the conductive layer INS-L1 in the first layer and the conductivelayer INS-L3 in the third layer, the present invention is not limited tothe configuration. As long as wiring patterns which are overlapped atleast partly in plan view can be formed, a wiring pattern in anarbitrary layer can be used as a wiring pattern as a component of thedelay element DLN.

Third Embodiment

FIG. 15 and FIGS. 16A to 16C are diagrams illustrating the structure ofa semiconductor device according to a third embodiment. In the first andsecond embodiments, the example of configuring the delay element DLN bythe wiring pattern in the interposer has been described. In the thirdembodiment, the wiring pattern as a component of the delay element DLNis formed in the logic semiconductor chip LCH-1. Since the delay elementDLN as a component of the digital filter 1002 is formed in the logicsemiconductor chip LCH-1, a semiconductor device using no interposer isdescribed as an example. Obviously, an interposer may be providedbetween the logic semiconductor chip LCH-1 and the package substratePPS-1.

FIG. 15 is a cross section of the semiconductor device according to thethird embodiment. FIGS. 16A to 16C are diagrams illustrating thestructure of the digital filter 1002. FIG. 16A is a plan view of thedigital filter 1002, and FIG. 16B is a cross section illustrating thestructure of the delay element DLN according to the third embodiment.

FIG. 15 is an A3-A3′ section in FIG. 16A. First, the semiconductordevice according to the third embodiment will be described withreference to FIG. 15. FIG. 15 is similar to FIG. 8. Specifically, theprint substrate PBS, the package substrate PPS-1, the ball SBL, and thebump SMB illustrated in FIG. 15 are the same as those in FIG. 8.Consequently, their description will not be repeated here.

A bump electrode LCH-PD3 is formed on the second main surface CHF2 ofthe logic semiconductor chip LCH-1. The bump electrode LCH-PD3 iscoupled to the bump electrode PPS-MPD formed on the first main surfacePPF1 of the package substrate by the bump SMB.

The logic semiconductor chip LCH-1 has the silicon substrate SSB and thewiring layer HSB formed on the main surface of the silicon substrateSSB. The wiring layer HSB has a plurality of conductive layers and aplurality of insulating layers which are alternately stacked. Althoughnot limited, description will be given on assumption that the wiringlayer HSB has three conductive layers. Obviously, the present inventionis not limited to the number of the layers.

To form the plurality of circuit blocks such as the transmission buffercircuit SCB1-1 and the reception buffer circuit RCB1-2, in the siliconsubstrate SSB, a plurality of semiconductor regions functioning as thesource and the drain of a field effect transistor (hereinbelow, calledMOS FET) are formed. The plurality of formed semiconductor regions arecoupled by the wiring pattern formed by the conductive layer in thewiring layer HCB. Consequently, the circuit blocks such as thetransmission buffer circuit SCB1-1 and the reception buffer circuitRCB1-2 are configured. In FIG. 15, the MOSFET is omitted and only a partsuch as the digital filter 1002 is illustrated.

In the logic semiconductor chip LCH-1 illustrated in FIG. 15,LCH-L10(R), LCH-L10(V), and LCH-L11(V) are wiring patterns formed by thefirst conductive layer LCH-L1 and disposed in the main surface of thesilicon substrate SSB. As will be described specifically later withreference to FIG. 16, the wiring pattern LCH-L10(R) is formed in thesilicon substrate SSB via an insulating layer (gate insulating field),and the wiring patterns LCH-L10(V) and LCH-L11(V) are formed so as to beohmic-coupled to the silicon substrate SSB. One end of the wiringpattern LCH-L10(R) is coupled to the wiring pattern LCH-L20(R) formed bythe second conductive layer LCH-L2 via the contact CT1, and the wiringpattern LCH-L20(R) is coupled to the wiring pattern LCH-L30(R) formed bythe third conductive layer LCH-L3 via the contact CT1.

The wiring pattern LCH-L30(R) is coupled to the bump electrode LCH-PD3via the contact CT1. The other end of the wiring pattern LCH-L10(R) iscoupled to the wiring pattern LCH-L21(V) formed by the second conductivelayer LCH-L2 via the contact CT1, and the wiring pattern LCH-L21(V) isfurther coupled to the wiring pattern LCH-L1(V) via the contact CT1.

The wiring pattern LCH-L10(V) is coupled to the wiring patternLCH-L20(V) formed by the second conductive layer LCH-L2 via the contactCT1, and the wiring pattern LCH-L20(V) is coupled to the wiring patternLCH-L30(V) formed by the third wiring layer LCH-L3 via the contact CT1.

FIG. 16A is a plan view including the section part illustrated in FIG.15. As illustrated in FIG. 16A, the wiring pattern LCH-L10(R) extends inthe lateral direction. In the third embodiment, as illustrated in FIG.16A, the wiring pattern LCH-L30(R) is coupled to the output terminal ofthe transmission buffer circuit SCB1-1 (FIG. 6). Consequently, atransmission signal from the transmission buffer circuit SCB1-1 isoutput to one end of the wiring pattern LCH-L10(R) and the bumpelectrode LCH-PD3. A signal according to the input signal is output fromthe one end of the wiring pattern LCH-L10(R) and combined with thetransmission signal output to the bump electrode LCH-PD3, and aresultant signal passes through a wiring pattern formed in the printsubstrate PBS and the like and is transmitted to the logic semiconductorchip LCH-2 (FIG. 6).

In FIG. 16A, two sets of the wiring patterns LCH-L10(V) and LCH-L20(V)are drawn. In FIG. 16B, only one set of the wiring patterns LCH-L10(V)and LCH-L20(V) on the right side in the two sets is drawn. The wiringpattern LCH-L30(V) is disposed like a mesh shape as illustrated in FIG.16B and is coupled to, for example, the transmission buffer circuitSCB1-1 and the reception buffer circuit RCB1-2. Those buffer circuitsoperate on, for example, the predetermined voltage Vs as a referencevoltage.

In the third embodiment, the delay element DLN is formed by the wiringpattern LCH-L10(R) and the silicon substrate SSB. In the thirdembodiment, in the equivalent circuit illustrated in FIG. 1B, not onlythe resistance R but also the conductance G can be increased and thesignal loss ratio can be controlled. By using the large dielectricconstant of the silicon substrate SSB, the delay amount per unit lengthof the wiring pattern LCH-L10(R) can be increased, and the size of thedelay element DLN can be made smaller.

Next, by using FIG. 16B, the delay element DLN according to the thirdembodiment will be described. The silicon substrate SSB is, for example,a P-type silicon substrate. For the wiring pattern LCH-L10(R), as aninsulating layer, a gate insulating film formed over the P-type siliconsubstrate is used. The other end of the wiring pattern LCH-L10(R) iscoupled to the P-type silicon substrate via the wiring patternsLCH-L11(V) and LCH-L21(V) which are ohmic-coupled to the P-type siliconsubstrate. The wiring pattern LCH-L10(R) operates as the gate electrodeof a MOSFET and, equivalent, a MOS diode (equivalent diode element)having the gate electrode to which the source or drain of the MOSFET iscoupled is formed. That is, a MOS diode of a distributed constant typeis formed and functions as the delay element DLN. The relativepermittivity of an oxide film used as an insulating layer in the siliconinterposer is about four. The relative permittivity of a resin materialused as an insulating layer in another interposer is about 3.1. Incomparison to the relative permittivity of those materials, the relativepermittivity of silicon is about 12. Consequently, delay per unit lengthof the delay element DLN can be made large, and the size of the delayelement DLN can be reduced. Since the equivalent MOS diode is used,current flows. The current corresponds to current passed by the parallelconductance G illustrated in FIG. 1B. As a result, the signal loss ratiocan be controlled not only by the resistance R illustrated in FIG. 1Bbut also the conductance G.

The above-described wiring patterns and the like are formed by the knownsemiconductor manufacturing technology. There is a case that theresistance value of the wiring pattern LCH-L10(R) extending in thelateral direction is too high. In this case, it is sufficient to use thestructure as illustrated in FIG. 16C as the structure of the delayelement DLN. Specifically, the wiring pattern LCH-L20(R) extends in thelateral direction and is coupled to the wiring pattern LCH-L21(V).Further, the extended wiring pattern LCH-L20(R) and the wiring patternLCH-L10(R) are coupled by a plurality of contacts CT1. In such a manner,the combined resistance of the wiring patterns LCH-L10(R) and LCH-L20(R)can be lowered.

In the third embodiment, the delay element DLN is formed in the logicsemiconductor chip LCH-1. Consequently, one end of the delay element DLNcan be disposed close to the output terminal of the transmission buffercircuit or the input terminal of the reception buffer circuit. By thearrangement, signal deterioration which occurs between the part of thewired-OR coupling (node WNR in FIG. 1) and the input terminal or theoutput terminal can be reduced.

In the third embodiment, the delay element DLN is equivalently formed bythe MOS diode, so that the resistance R relative to the predeterminedvoltage Vs (ground voltage of the circuit) per unit length of the signalwire and the voltage wire as components of the delay element DLN can bemade smaller than the resistance R relative to the predetermined voltageVs per unit length of the signal line 1000. In other words, theconductance G relative to the predetermined voltage Vs (ground voltageof the circuit) can be increased.

Although the P-type silicon substrate has been described as an exampleof the silicon substrate, an N-type silicon substrate may be used. Thesilicon substrate illustrated in FIGS. 16B and 16C may be a well regionof the P type or the N type.

Fourth Embodiment

FIG. 17 and FIGS. 18A to 18C are diagrams illustrating the structure ofa semiconductor device according to a fourth embodiment. Also in thefourth embodiment, like in the third embodiment, the delay element DLNis formed in the logic semiconductor chip LCH-1. FIG. 17 and FIGS. 18Ato 18C are similar to FIG. 15 and FIGS. 16A to 16C and the differentpoints will be mainly described. In the third embodiment described withreference to FIG. 15 and FIGS. 16A to 16C, the delay element DLN iscomprised of the MOS diode of the distributed constant type. On theother hand, in the fourth embodiment, the delay element DLN is comprisedof a PN junction diode (equivalent diode element) of the distributedconstant type.

FIG. 17 is a cross section illustrating a section of a semiconductordevice like FIG. 15. FIGS. 18A to 18C are diagrams illustrating thestructure of the delay element like FIGS. 16A to 16C. FIG. 18A is a planview illustrating the structure of the delay element, and the A4-A4′section in FIG. 18A is illustrated in FIG. 17. FIG. 18B is a crosssection illustrating a section of the delay element like FIG. 16B.

In the fourth embodiment, to form the PN junction diode of thedistributed constant type, in the silicon semiconductor substrate SSB, asemiconductor region of the conduction type opposite to that of thesilicon substrate SSB is formed. In FIGS. 17 and 18B, SSB-n indicates asemiconductor region formed in the silicon substrate SSB. The siliconsubstrate SSB is, for example, a P-type silicon substrate. In this case,the semiconductor region SB-n is an N-type semiconductor region of theconduction type opposite to the P type. Consequently, in the junctionpart between the P-type silicon substrate SSB and the N-typesemiconductor region, a PN-type diode is formed.

The N-type semiconductor region SSB-n extends in the lateral directionin FIG. 17 and FIGS. 18B and 18C. The N-type semiconductor region SSB-nextending in the lateral direction is ohmic-coupled to the wiringpattern LCH-L10(R). In the third embodiment, the wiring patternLCH-L10(R) is formed over the silicon substrate SSB via the gate oxidefilm and functions as the gate electrode. In the fourth embodiment, thewiring pattern LCH-L10(R) functions as the electrode of the PN junctiondiode.

In this case, the P-type silicon substrate SSB is coupled to thepredetermined voltage Vs (for example, the ground voltage of thecircuit). In the embodiment, since the delay element DLN is comprised ofthe PN junction diode, current flowing in the PN junction diode can beincreased. An equivalent circuit of the delay element DLN comprised ofthe PN junction diode is the same as the equivalent circuit illustratedin FIG. 1B. Since diode current flowing in the PN junction diode can beregarded as current flowing in the PN junction diode which is coupled inparallel in the equivalent circuit, according to the fourth embodiment,the delay element DLN having larger conductance G can be formed.

FIG. 18C is similar to FIG. 16C. In FIG. 18C, the wiring patternLCH-L20(R) is disposed so as to extend along the wiring patternLCH-L10(R). The extended wiring pattern LCH-L20(R) is coupled inparallel to the wiring pattern LCH-L10(R) by the plurality of contactsCT1. In such a manner, the combined resistance of the wiring patternsLCH-L10(R) and LCH-L20(R) can be reduced. That is, the resistance of theelectrode of the PN junction diode can be reduced. In the fourthembodiment, to reduce the resistance of the electrode of the diode, thewiring pattern LCH-L20(R) is extended. Consequently, different from thethird embodiment, the extended wiring pattern LCH-L20(R) is electricallyisolated from the wiring pattern LCH-L21(V) coupled to the P-typesilicon substrate.

Although the case that the silicon substrate SSB is the P-type siliconsubstrate has been described as an example, like the third embodiment,the present invention is not limited to the case. The silicon substrateSSB may be an N-type silicon substrate or a well of the P type or the Ntype. Obviously, in the case of the N-type silicon substrate or the wellof the N type, a P-type semiconductor region is used in place of theN-type semiconductor region SSB-n.

In the third and fourth embodiments, the digital filter coupled to theoutput terminal of the transmission buffer circuit SCB1-1 has beendescribed as an example. Obviously, the digital filter described in thethird and fourth embodiments may be coupled to the input terminal of thereception buffer circuit RCB1-2.

Fifth Embodiment

In the first to fourth embodiments, the signal line transmitting asingle-phase signal has been described as an example. The digital filtercan be also applied to a signal line transmitting a differential signal.Also in this case, while reducing the consumption power, a signal can berestored (shaped). A mode of using the digital filter for the signalline transferring the differential signal will be described as a fifthembodiment.

FIG. 19 is a block diagram illustrating the configuration of a digitalfilter according to a fifth embodiment. In FIG. 19, 1001P denotes atransmission buffer circuit having a pair of output terminals, and 1003Pindicates a reception buffer circuit having a pair of input terminals.The transmission buffer circuit 1001P is provided, for example, for theSerDes circuit in the logic semiconductor chip LCH-2 illustrated in FIG.6, and the reception buffer circuit 1003P is provided for the SerDescircuit in the logic semiconductor chip LCH-1 illustrated in FIG. 6. Thetransmission buffer circuit 1001P receives a signal to be transmittedand generates differential signals according to the received signal. Thetransmission buffer circuit 1001P generates, for example, a pair ofserial signals whose phases are inverted as differential signals attransfer speed of 56 Gpbs. The generated differential signals aresupplied from the pair of output terminals of the transmission buffercircuit 1001P to ends SNI1 and SNI2 of a pair of signal lines (first andsecond signal lines) 1000P1 and 1000P2. Specifically, one of the pair ofserial signals generated is supplied to the end ENI1 of the signal line(first or second signal line) 1000P1, and the other serial signal issupplied to the end SNI2 of the signal line (second or first signalline) 1000P2.

The pair of signal lines 1000P1 and 1000P2 is comprised of wiringpatterns and contacts coupling the transmission buffer circuit 1001Pdisposed in the logic semiconductor chip LCH-1 and the reception buffercircuit 1003P disposed on the logic semiconductor chip LCH-2. Forexample, a pair of wiring patterns and the like formed in the printsubstrate PBS are included in the pair of signal lines 1000P1 and1000P2.

The differential signals supplied to the ends SNI1 and SNI2 of the pairof signal lines 1000P1 and 1000P2 propagate through the signal lines1000P1 and 1000P2 and are transmitted to ends SNO1 and SNO2 of thesignal lines 1000P1 and 1000P2. The pair of signal lines 1000P1 and1000P2 are coupled to a pair of digital filters 1002P1 and 1002P2 atnodes WRN1 and WRN2, respectively.

A pair of input terminals of the reception buffer circuit 1003P iscoupled to the nodes WRN1 and WRN2. Specifically, one of the pair ofinput terminals of the reception buffer circuit 1003P is coupled to thenode WRN1, and the other input terminal is coupled to the node WRN2. Thereception buffer circuit 1003P amplifies the differential signalssupplied to the pair of input terminals and outputs a resultant signal.It can be regarded that the reception buffer circuit 1003P has adifferential circuit coupled to the pair of input terminals. In thiscase, the differential signals supplied to the pair of input terminalsare amplified by the differential circuit. An output from the receptionbuffer circuit 1003P is processed in the SerDes circuit.

The pair of digital filters 1002P1 and 1002P2 have the sameconfiguration which is the same as that of the digital filter 1002described in the first to fourth embodiments. The digital filter 1002P1has a delay element DLN1 having a pair of ends DN1 and DN2, and thedigital filter 1002P2 has a delay element DLN2 having a pair of ends DN1and DN2.

At node WRN, the one end DN2 of the delay element DLN1 is wired-ORcoupled to the end SNO1 of the signal line 1000P1 and one of the inputterminals of the reception buffer circuit 1003P and the other end DN1 iscoupled to the predetermined voltage Vs. Also in the delay element DLN2,like the delay element DLN1, at the node WRN, one end DN2 is wired-ORcoupled to the end SNO2 of the signal line 1000P2 and the other inputterminal of the reception buffer circuit 1003P, and the other end DN1 iscoupled to the predetermined voltage Vs.

Correspondence between the digital filter 1002 described in the firstembodiment and the pair of digital filters 1002P1 and 1002P2 will bedescribed as follows. Each of the delay elements DLN1 and DLN2illustrated in FIG. 19 corresponds to the delay element DLN described inFIG. 1A and operates in a manner similar to the delay element DLN.

Since the operation is similar, detailed description will not berepeated. One end DN2 of each of the delay elements DLN1 and DLN2functions as a signal input/output terminal. A signal according to thesignal distribution ratio “b” in signals output from the end SNO1 of thesignal line 1000P1 is input to the delay element DLN1 and, after theround-trip signal delay UT/m, an attenuated signal is output (solid-linearrow). The signal output from the delay element DLN1 is combined at thewired-OR-coupling part (node WRN1), and a combined signal is supplied toone of the input terminals of the reception buffer circuit 1003P.Similarly, a signal according to the signal distribution ratio “b” insignals output from the end SNO2 of the signal line 1000P2 is input tothe delay element DLN2 and, after the round-trip signal delay UT/m, anattenuated signal is output (alternate long and short dash arrow). Thesignal output from the delay element DLN2 is combined at thewired-OR-coupling part (node WRN2), and a combined signal is supplied tothe other input terminal of the reception buffer circuit 1003P.

As described above, to each of the pair of input terminals of thereception buffer circuit 1003P, the signal restored (shaped) by thecombination is supplied. Since the pair of digital filters 1002P1 and1002P2 is formed by the delay elements DLN1 and DLN2 as passiveelements, respectively, while reducing consumption power, signaldeterioration caused by the loss in the pair of signal lines can berestored.

Each of the delay elements DLN1 and DLN2 may be formed in the interposeras described in the first and second embodiments or formed in thesemiconductor chip as described in the third and fourth embodiments.FIG. 19 illustrates the example of coupling the pair of digital filters1002P1 and 1002P2 to the pair of input terminal sides of the receptionbuffer circuit 1003P. The pair of digital filters 1002P1 and 1002P2 maybe coupled to the pair of output terminal sides of the transmissionbuffer circuit 1001P.

Sixth Embodiment

FIG. 20 is a block diagram illustrating the configuration of a digitalfilter according to a sixth embodiment. Also in the sixth embodiment, adigital filter adapted to a signal line transmitting differentialsignals is provided. Since FIG. 20 is similar to FIG. 19, the differentpoints will be mainly described.

In FIG. 20, the pair of signal lines 1000P1 and 1000P2, the transmissionbuffer circuit 1001P, and the reception buffer circuit 1003P are thesame as those of FIG. 19, so that the description will not be repeated.In FIG. 19, the digital filters 1002P1 and 1002P2 are provided for thesignal lines 1000P1 and 1000P2, respectively. In contrast, in the sixthembodiment, one digital filter 1002P is provided for the pair of signallines 1002P1 and 1002P2.

The digital filter 1002P includes the delay element DLN having a pair ofends DNIO1 and DNIO2. The end DNIO1 as one of the ends of the delayelement DLN is wired-OR-coupled to the end SNO1 of the signal line1001P1 and one input terminal of the reception buffer circuit 1003P atthe node WRN1. Similarly, the other end DNIO2 of the delay element DLNis wired-OR-coupled to the end SNO2 of the signal line 1001P2 and theother input terminal of the reception buffer circuit 1003P at the nodeWRN2. Different from the delay element DLN described in the first tofifth embodiments, the delay element DLN does not have an end coupled tothe predetermined voltage Vs.

In FIG. 20, the delay element DLN is drawn folded in its plane shape.However, the invention is not limited to the plane shape. For example,like the wiring pattern INS-L10 illustrated in FIG. 7, the plane shapemay be linear. In the case of using the wiring pattern INS-L10 as thesignal wire in the delay element DLN illustrated in FIG. 20, the wiringpatterns INS-L10 and INS-L13 are separated, one end of the wiringpattern INS-L10 is coupled to the node WRN1 in FIG. 20, and the otherend is coupled to the node wRN2 in FIG. 20. Also in this case, in planview, in a manner similar to FIG. 7, the wiring pattern L10 extendsbetween the wiring patterns INS-L11 and INS-L12 as voltage wires ascomponents of the delay element DLN in parallel to the wiring patternsINS-L11 and INS-L12.

The delay element DLN illustrated in FIG. 20 is set to have a round-tripsignal delay corresponding to sum of a round-trip signal delay of thedelay element DLN1 and a round-trip signal delay of the delay elementDLN2. The signal loss amount of the delay element DLN illustrated inFIG. 20 is set to be the sum of a signal loss amount (db) of the delayelement DLN1 and a signal loss amount (db) of the delay element DLN2.Since the delay elements DLN1 and DLN2 illustrated in FIG. 19 have thesame configuration, the delay element DLN illustrated in FIG. 20 isformed by a wiring pattern having a length which is twice as long asthat of the delay element DLN1.

A pair of signals (serial signals) forming differential signals can beconsidered that, when viewed from one signal (other signal), the othersignal (one signal) functions as a reference potential. That is, thephases function as reference potentials each other. In the sixthembodiment, one of differential signals is input to the end DNIO1 of thedelay element DLN, and the other differential signal is input to theother terminal DNIO2 of the delay element DLN. Consequently, when thesignal input to the end DNIO1 is viewed, the reference potential to thesignal is supplied to the other end DNIO2. Similarly, when the signalinput to the other end DNIO2 is viewed, the reference potential to thesignal is supplied to the end DNIO1.

As a result, the signal input to the end DNIO1 of the delay element DLNis reflected by the other end DNIO2 of the delay element DLN, and thereflected signal is output from the end DNIO1 (solid-line arrow).Similarly, the signal input to the other end DNIO2 of the delay elementDLN is reflected by the end DNIO1 of the delay element DLN, and thereflected signal is output from the other end DNIO2 (arrow of thealternate long and short dash line).

Since the pair of signals input to one end DNIO1 and the other end DNIO2of the delay element DLN are differential signals, when the signal inputto the other end DNIO2 (or one end DNIO1) is used as a referencepotential, the signal input to the end DNIO1 (or the other end DNIO2)has a value twice as high as the center potential of the pair of signals(predetermined voltage Vs). Consequently, the signal attenuation amountof the delay element DLN is set to twice (sum of the signal attenuationamount (db) of the delay element DLN1 and the signal attenuation amount(db) of the delay element DLN2).

The pair of signals as differential signals change at the samefrequency. Consequently, when the signal input to the other end DNIO2functions as the reference potential, in the case of regarding thereference potential as a fixed reference potential which does not changewith time, it can be regarded that the signal input to the end DNIO1 hasfrequency of twice. Consequently, the round-trip signal delay of thedelay element DLN is doubled.

As the delay element DLN is sufficiently short, the round-trip signaldelay of the delay element DLN depends on the length of the delayelement DLN. Therefore, by setting the sum of the delay elements DLN1and DLN2 illustrated in FIG. 19 to the length of the delay element DLN,a round-trip signal delay of twice can be obtained.

The round-trip signal delay is a value obtained by dividing the datawidth interval UT by the integer “m”, and the round-trip signal delay oftwice relates to the case where the integer “m” is 1. “Twice” is anexample, and it is sufficient to determine the length or the like of thedelay element DLN so that the round-trip signal delay occurs integer (m)number of times including one in the one data width interval UT.

A signal output from the end DNIO1 of the delay element DLN is combinedat the node WRN1. Similarly, a signal output from the end DNIO2 of thedelay element DLN is combined at the node WRN2. As a result, therestored (shaped) differential signals are supplied to the pair of inputterminals of the reception buffer circuit 1003P. That is, while reducingconsumption power, the waveform deformed in the signal line can beshaped.

In the sixth embodiment, when noise of the same phase occurs in the pairof signal lines 1000P1 and 1000P2, the reference potential supplied tothe end DNIO2 or DNIO1 of the delay element changes according to thenoise. As a result, an output signal from the end DNIO1 or DNIO2 is notinfluenced by the noise. That is, the influence of the noise can bereduced.

Although FIG. 20 illustrates the example that the digital filter 1002Pis provided on the reception buffer circuit 1003P side, the digitalfilter 1002P may be provided on the transmission buffer circuit 1001Pside. As described in the first to fourth embodiments, the delay elementDLN may be formed in the interposer or the semiconductor chip.

Seventh Embodiment

In the fifth and sixth embodiments, the digital filter used for thesignal lines (differential signal lines) transmitting differentialsignals has been described. In a seventh embodiment, a more concretemode of a digital filter used for the signal lines (differential signallines) transmitting differential signals is provided.

Differential Signal and Common Mode Signal

First, an example of differential signals transmitted through the signallines will be described with reference to FIG. 24. FIG. 24 is a waveformchart illustrating voltage waveforms of differential signals. In thediagram, the horizontal axis indicates time “t” and the vertical axisindicates voltage. In the vertical axis, Vref denotes reference voltage.+Vd indicates voltage whose voltage value is higher than the referencevoltage Vref, and −Vd indicates voltage lower than the reference voltageVref.

The differential signals are a pair of signals whose voltages changecomplementarily with lapse of time. In FIG. 24, the pair of signals areindicated as reference characters SSI and/SSI. When the signal SSIchanges with lapse of time in the range of a voltage higher than thereference voltage Vref, the signal/SSI changes with lapse of time in therange of a voltage lower than the reference voltage Vref. Changes of thesignals SSI and/SSI accompanying the lapse of time are complementarily.Specifically, when the voltage of the signal SSI changes so as toincrease from the reference voltage Vref toward the voltage +Vd, thevoltage of the signal/SSI changes so as to decreases from the referencevoltage Vref toward the voltage −Vd. Similarly, when the signal SSIchanges so as to decrease from the voltage +Vd toward the referencevoltage Vref, the voltage of the signal/SSI changes so as to increasefrom the voltage −Vd toward the reference voltage Vref.

For example, the differential circuit detects and/or amplifies thevoltage difference between the pair of signals SSI and/SSI. In thespecification, hereinafter, when there is no need to clearly describe,each of the pair of signals SSI and/SSI will be called a differentialsignal.

In FIG. 24, +Vpd indicates peak voltage when the differential signal SSIchanges, and −Vpd indicates peak voltage when the differentialsignal/SSI changes. Description will be given on assumption that theabsolute value of the voltage of the differential signal SSI and theabsolute value of the voltage of the differential signal/SSI are equal.Consequently, the absolute value of the voltage from the referencevoltage Vref to the peak voltage +Vpd and the absolute value of thevoltage from the reference voltage Vref to the peak voltage −Vpd areequal.

In FIG. 24, CMI indicates a common mode signal synchronized with thedifferential signals SSI and/SSI. In FIG. 24, the common mode signal CMIis indicated as a signal having the same phase as that of thedifferential signal SSI. Specifically, the common mode signal CMIincreases synchronously with increase in the differential signal SSI anddecreases synchronously with decrease in the differential signal SSI. InFIG. 24, the peak voltage of the common mode signal CMI relative to thereference voltage Vref is indicated as +Vpc.

When the common mode signal CMI is superimposed on each of thedifferential signals SSI and/SSI as noise, the voltage of each of thedifferential signals SSI and/SSI increases only by the voltage amount ofthe common mode signal CMI. In this case, since the common mode signalCMI is superimposed as noise, the common mode signal CMI can be regardedas common mode noise CMI and therefore will be also called as commonmode noise CMI.

Even when the voltages of the differential signals SSI and/SSI changedue to the common mode noise CMI, the voltage difference does notchange, so that the differential circuit can output a detection resultand/or an amplification result which is not influenced by the commonmode noise CMI.

Single-End Digital Filter

In the fifth embodiment, as illustrated in FIG. 19, the pair of signals(differential signals SSI and/SSI) are supplied from the pair of outputterminals of the transmission buffer circuit 1001P to the ends SNI1 andSNI2 of the pair of signal lines 1000P1 and 1000P2. The differentialsignals SSI and/SSI propagate through the signal lines 1000P1 and 1000P2and are transmitted to the ends SNO1 and SNO2 of the signal lines 1000P1and 1000P2. In the fifth embodiment, the pair of signal lines 1000P1 and1000P2 are coupled to the pair of digital filters 1002P1 and 1002P2 atthe nodes WRN1 and WRN2.

As illustrated in FIG. 19, the digital filters 1002P1 and 1002P2 havethe delay elements DLN1 and DLN2 each having the other end DN1 coupledto the predetermined voltage Vs. Since the other end DN1 is coupled tothe predetermined voltage Vs such as the ground voltage, each of thedigital filters 1002P1 and 1002P2 illustrated in FIG. 19 can be regardedas the single-end digital filter.

Common Mode Signal

The transmission buffer circuit 1001P illustrated in FIG. 19 supplies apair of differential signals to the pair of signal lines 1000P1 and1000P2. In reality, the transmission buffer circuit 1001P generates somecommon mode signals. When an asymmetric part exists between the signallines 1000P1 and 1000P2, there is a case that a common mode signal isgenerated. The common mode signal generated in such a manner is appliedas noise to each of the signal lines 1000P1 and 1000P2. The digitalfilters (single-end digital filters) 1002P1 and 1002P2 illustrated inFIG. 19 operate so as to equalize the given common mode signal (commonmode noise). The inventors of the present invention examined that, inthe case of the single-end digital filter, not only the differentialsignals but also the common mode noise are equalized. The result of theexamination will now be described.

FIGS. 25A to 25C are diagrams illustrating eye patterns in the case ofconfiguring each of the digital filters 1002P1 and 1002P2 by asingle-end digital filter. In FIGS. 25A to 25C, the horizontal axisindicates time, and the vertical axis indicates signal voltage. The eyepattern is obtained under conditions similar to those of FIGS. 9A and9B. Specifically, FIGS. 25A to 25C illustrate a case that an FR4substrate is used as the print substrate, the logic semiconductor chipsLCH-1 and LCH-2 as illustrated in FIG. 6 are mounted on the printsubstrate so that the interval becomes four inches, and differentialsignals in the NRZ format are supplied from the logic semiconductor chipLCH-2 to the logic semiconductor chip LCH-1 at transmission speed of 56Gbps via the wiring pattern of the print substrate. As the interposer, asilicon interposer is used. In FIGS. 25A to 25C, the reception buffercircuit 1003P as illustrated in FIG. 19 is used as the reception buffercircuit of the logic semiconductor chip LCH-1, and a pair ofdifferential signals are supplied to the reception buffer circuit 1003Pvia the pair of signal lines 1000P1 and 1000P2 as illustrated in FIG.19.

FIG. 25A is a diagram of eye patterns drawn by overlapping signalvoltage waveforms at the input terminal of the reception buffer circuit1003P in a state where the digital filters 1002P1 and 1002P2 are coupledto the nodes WRN1 and WRN2 as illustrated in FIG. 19. In FIG. 25A, adotted waveform SSIB is a signal voltage waveform generated byequalizing components of the differential signals SSI and/SSI, and asolid-line waveform CMIB indicates a signal voltage waveform generatedby equalizing components of the common mode noise CMI. FIG. 25illustrates the case where the value of the peak voltage +Vpc (FIG. 24)of the common mode noise CMI is 20% of the voltage difference betweenthe peak voltage +Vpd of the differential signal SSI and the peakvoltage −Vpd of the differential signal/SSI.

FIG. 25B is a diagram illustrating eye patterns obtained by extractingthe signal voltage waveform SSIB generated by equalizing the componentsof the differential signals SSI and/SSI from FIG. 25A. FIG. 25C is adiagram illustrating eye patterns obtained by extracting the signalvoltage waveform CMIB generated by equalizing the components of thecommon mode signal CMI from FIG. 25A.

It is understood from FIG. 25B that since the eye pattern is open, thecomponents of the differential signals SSI and/SSI are equalized by thedigital filters 1002P1 and 1002P2 and the transmitted differentialsignals can be identified. However, as illustrated in FIG. 25C, the eyepattern is open with respect to the components of the common mode signalCMI. That is, the components of the common mode signal CMI are alsoequalized by the digital filters 1002P1 and 1002P2. As a result, asillustrated in FIG. 25A, in the eye pattern, the signal voltage waveformSSBIB generated by equalizing the components of the differential signalsSSI and/SSI and the signal voltage waveform CMIB generated by equalizingthe components of the common mode noise mixedly exist.

In the mixedly existing state, for example, in the case of identifyingthe logic value of a transmitted differential signal (serial signal) inthe reception buffer circuit 1003P, the logic value of the differentialsignal is erroneously identified on the basis of the signal voltagewaveform CMIB of the common mode signal CMI, and it causes erroneousoperation.

Skew

It is desirable that the pair of differential signals SSI and/SSI changesynchronously with each other but there is a case that delay (skew)occurs between the differential signals SSI and/SSI. For example, delayoccurs between the differential signals SSI and/SSI due to variations inthe characteristic of the transmission buffer circuit 1001P (FIG. 19),variations in delay between the signal lines 1000P1 and 1000P2, and thelike. As an example, a timing the voltage of the differential signal/SSIchanges delays from a timing the voltage of the differential signal SSIchanges.

The inventors of the present invention also examined the eye patterns inthe case where there is a skew between the pair of differential signalsSSI and/SSI. FIGS. 26A to 26C are diagrams illustrating eye patternsexamined by the inventors of the present invention. FIGS. 26A to 26Calso illustrate the eye patterns in the case where each of the digitalfilters 1002P1 and 1002P2 is a single-end digital filter. FIGS. 26A to26C illustrate eye patterns in the case where skew of three picoseconds(psec) exists between the differential signals SSI and/SSI due to delayvariations in the signal lines. Also in FIGS. 26A to 26C, the horizontalaxis indicates time, and the vertical axis indicates signal voltage. Theeye patterns illustrated in FIGS. 26A to 26C are obtained by conditionssimilar to those of FIGS. 25A to 25C.

In FIG. 26A, the dotted waveform SSIB indicates the signal voltagewaveform generated by equalizing the components of the differentialsignals SSI and/SSI, and the solid-line waveform SKYB indicates a signalvoltage waveform generated by equalizing the skew. Like FIG. 25B, FIG.26B is a diagram illustrating eye patterns obtained by extracting thesignal voltage waveform SSIB generated by the components of thedifferential signals SSI and/SSI from FIG. 26A. FIG. 26C is a diagramillustrating the eye patterns by extracting the signal voltage waveformSKYB generated by the skew.

FIGS. 25A to 25C illustrate the case where the common mode noise CMI ofthe phase same as that of the differential signals is mixed in thedifferential signals SSI and/SSI. Consequently, the phase of the signalvoltage waveform SSIB generated by the components of the differentialsignals and that of the signal voltage waveform CMIB generated by thecomponent of the common mode noise CI are the same. That is, the phaseof the eye pattern by the signal voltage waveform SSIB and that of theeye pattern by the signal voltage waveform CMIB are the same, and thetiming the eye pattern by the signal voltage waveform SSIB is open andthat the eye pattern by the signal voltage waveform CMIB is open match.In contrast, in FIGS. 26A to 26C, the timing the eye pattern by thesignal voltage waveform SKYB caused by skew is open is deviated from thetiming the eye pattern generated by the signal voltage waveform SSIB isopen.

Also in the case where the skew exists, as illustrated in FIG. 26C, theeye pattern according to the skew exists, so that an erroneous operationis caused like the case where the common mode signal CMI is mixed.

Configuration of Digital Filter

FIGS. 27A and 27B are diagrams illustrating the configuration of adigital filter according to the seventh embodiment and an equivalentcircuit of the digital filter. FIG. 27A illustrates the configuration ofthe digital filter, and FIG. 27B illustrates an equivalent circuit ofthe digital filter depicted in FIG. 27A.

Since FIG. 27A is similar to the configuration of the digital filterillustrated in FIG. 20, only the different point will be described. Inthe digital filter illustrated in FIG. 20, the delay element DLN iscomprised of a bended delay wire (hereinbelow, also called delay line).On the other hand, in the seventh embodiment, the delay element DLN iscomprised of a delay line of a linear shape. However, also in theseventh embodiment, as illustrated in FIG. 20, the delay element DLN maybe also comprised of a bended delay line. Except for the point that thedelay element DLN is comprised of a delay line of a linear shape, theconfiguration and operation of the digital filter illustrated in FIG.27A are similar to those of FIG. 20.

The equivalent circuit illustrated in FIG. 27B is similar to thatillustrated in FIG. 1B. The different point is that the ends DNIO1 andDNIO2 of the delay element DLN are coupled to the signal lines 1000P1and 1000P2. That is, the end DNIO1 of the delay element DLN iswired-OR-coupled to the signal line 1000P1, and the end DNIO2 iswired-OR-coupled to the signal line 1000P2. Each of the signal lines1000P1 and 1000P2 is coupled between the transmission buffer circuit1001P and the reception buffer circuit 1003P, and the wired-OR couplingto the delay element DLN is performed near the reception buffer circuit1003P.

In the equivalent circuit diagram illustrated in FIG. 27B, like FIG. 1B,the inductance L, the delay element DLN is expressed by a π-typedistributed constant circuit comprised of the inductance L, theresistance R, the capacitance C, and the conductance G. Although thedistributed constant circuit is expressed so that the inductance L andthe resistance R are coupled to each of the ends DNIO1 and DNIO2 in FIG.27B, the invention is not limited to the configuration. As will bedescribed later, the voltage wire is disposed along the delay line as acomponent of the delay element DLN. To the voltage wire, thepredetermined voltage Vs is supplied. The predetermined voltage Vssupplied to the voltage wire functions as the voltage Vs illustrated inFIG. 27B. The equivalent circuit illustrated in FIG. 27B will bedescribed that a capacitance and a conductance formed in parallelbetween the voltage wire and the delay wire are expressed by thecapacitance C and the conductance G configuring the distributed constantcircuit illustrated in FIG. 27B.

In a manner similar to the six embodiment, the digital filter 1002Pincludes the delay element DLN having the pair of ends DNIO1 and DNIO2.The end DNIO1 of the delay element DLN is wired-OR-coupled to the endSNO1 of the signal line 1001P1 and one of the input terminals of thereception buffer circuit 1003P at the node WRN1. The other end DNIO2 ofthe delay element DLN is wired-OR-coupled to the end SNO2 of the signalline 1001P2 and the other input terminal of the of the reception buffercircuit 1003P at the node WRN2. As described, the delay element DLN doesnot have an end coupled to the predetermined voltage Vs in a mannersimilar to the sixth embodiment.

As understood from the description of FIG. 20, when the differentialsignal SSI (or the other differentia signal/SSI) is viewed as one of thepair of differential signals SSI and/SSI, the other differentialsignal/SSI (or the differential signal SSI) can function as thereference potential. That is, one of the differential signals or theother differential signal functions as the reference potential. In FIG.27A, in the case where the differential signal SSI as one of thedifferential signals is input to one end DNIO1 of the delay element DLNand the other differential signal/SSI is input to the other end DNIO2 ofthe delay element DLN, when the differential signal input to the endDNIO1 is viewed, the reference potential to the differential signal SSIis supplied to the other end DNIO2. Similarly, when the differentialsignal/SSI input to the other end DNIO2 is viewed, the referencepotential to the differential signal/SSI is supplied to the end DNIO1.

As a result, the differential signal SSI input to the end DNIO1 of thedelay element DLN is reflected by the other end DNIO2 of the delayelement DLN, and the reflected differential signal is output from theend DNIO1. Similarly, the differential signal/SSI input to the other endDNIO2 of the delay element DLN is reflected by the end DNIO1 of thedelay element DLN, and the reflected signal is output from the other endDNIO2.

The differential signal output from the end DNIO1 of the delay elementDLN is combined with the signal component (1-b) of the differentialsignal SSI from the signal line 1000P1 at the node WRN1. Similarly, thedifferential signal output from the end DNIO2 of the delay element DLNis combined with the signal component (1-b) of the differentialsignal/SSI from the signal line 1000P2 at the node WRN2. As a result,the restored (shaped) differential signals SSI and/SSI are supplied tothe pair of input terminals of the reception buffer circuit 1003P. Thatis, while reducing consumption power, the waveforms of the differentialsignal components deteriorated in the signal lines can be shaped.

In the seventh embodiment, when the common mode noise enters the pair ofsignal lines 1000P1 and 1000P2, the potentials of the ends DNIO2 andDNIO1 of the delay element change similarly according to the common modenoise. That is, according to the common node noise, the potentials inthe ends DNIO1 and DNIO2 change with the same phase. Consequently, whenthe potential at one of the ends DNIO1 and DNIO2 is regarded as thereference potential, the potential at the other end is not influenced bythe common mode noise. As a result, the components of the differentialsignals in the signals supplied to the ends DNIO1 and DNIO2 areequalized by the delay element DLN. On the other hand, a change in thepotentials of the same phase in the ends DNIO1 and DNIO2 caused by thecommon mode noise is transmitted to the reception buffer circuit 1003Pwithout being equalized by the delay element DLN. In other words, thedigital filter 1002P according to the seventh embodiment outputs thedeterioration of the waveform caused by the components of the commonmode noise without executing the function of equalization.

It can prevent formation of the opened eye pattern by the signal voltagewaveform CMIB caused by the common mode noise described with referenceto FIG. 25. As a result, erroneous operation of erroneously specifyingthe common mode noise CMI as a transmitted differential signal can beprevented. Although the common mode noise CMI is transmitted as apotential change of the same phase to the reception buffer circuit1003P, the reception buffer circuit 1003P has the differential circuit.Consequently, even when the potential changes of the same phase aresupplied, it does not exert an influence on detection and/oramplification.

Also in the case where a skew exists between the differential signalsSSI and/SSI, when the differential signals SSI and/SSI change in thesame voltage direction by a skew, the digital filter 1002P executes anoperation similar to that for the common mode noise. That is, when bothof the differential signals SSI and/SSI change in the same voltagedirection, that is, the direction of the voltage +Vd illustrated in FIG.24 by the skew, both of the ends DNIO1 and DNIO2 of the delay elementDLN change to the direction of the voltage +Vd. Consequently, like thecase of the common mode noise, the digital filter 1002P transmits thechange of the voltage to the reception buffer circuit 1003P withoutperforming equalization.

When a skew exits and the differential signals SSI and/SSI becomevoltages similar to the common mode noise, equalization by the delayelement is not performed. Consequently, opening of the eye pattern bythe signal voltage waveform SKYB generated by the skew can besuppressed. As a result, occurrence of erroneous operation caused by theskew can be reduced.

Eye Pattern

FIGS. 28A to 28C and FIGS. 29A to 29C illustrate eye patterns in thecase of wired-OR-coupling the delay element DLN as a component of thedigital filter 1002P to each of the signal lines 1000P1 and 1000P2 nearthe reception buffer circuit 1003P.

Common Mode Noise

The eye patterns illustrated in FIGS. 28A to 28C are obtained by thesame conditions as those described with reference to FIGS. 25A to 25C.FIGS. 28A to 28C illustrate the case that the common mode signal CMI issupplied to the signal lines 1000P1 and 1000P2. The peak voltage +Vpc(FIG. 24) of the common mode signal CMI supplied to the signal lines1000P1 and 1000P2 has the same value as that described in FIGS. 25A to25C, and the case where the voltage difference between the peak voltage+Vpd of the differential signal SSI and the peak voltage −Vpd of thedifferential signal/SSI is 20%.

In FIGS. 28A to 28C, the horizontal axis indicates time, and thevertical axis indicates voltage. FIG. 28A illustrates the eye patternsby the signal voltage waveform at the input terminal of the receptionbuffer circuit 1003P. In FIG. 28A, the dotted waveform SSIB is a signalvoltage waveform generated by equalizing components of the differentialsignals SSI and/SSI. In FIG. 28A, the solid-line waveform CMIB indicatesa signal voltage waveform generated by equalizing components of thecommon mode signal CMI. As illustrated in FIG. 28A, the eye patternformed by the signal voltage waveform SSIB is open. On the other hand,it is difficult to recognize an eye pattern of the signal voltagewaveform CMIB.

FIG. 28B is a diagram illustrating eye patterns obtained by extractingthe signal voltage waveform SSIB generated by equalizing the componentsof the differential signals SSI and/SSI from FIG. 28A. FIG. 28C is adiagram illustrating eye patterns obtained by extracting the signalvoltage waveform CMIB from FIG. 28A.

The components of the differential signals SSI and/SSI are equalized bythe digital filter 1002P and the deformed waveform is shaped.Consequently, as illustrated in FIGS. 28A and 28B, the eye patternsformed by the signal voltage waveform SSIB are open, and logic valuesexpressed by the differential signals SSI and/SSI can be identified withhigh precision and specified.

On the other hand, the components of the common mode signal CMI are notequalized by the digital filter 1002P and supplied as they are to theinput terminal of the reception buffer circuit 1003P. Consequently, asillustrated in FIGS. 28A and 28C, it is difficult to identify the eyepattern, and the components of the common mode signal CMI can beprevented from being erroneously identified as a logic value of adifferential signal which is transmitted.

Skew

The eye patterns illustrated in FIGS. 29A to 29C are obtained under thesame conditions as those described in FIGS. 25A to 25C. FIGS. 29A to 29Cillustrate the case where a skew of three picoseconds exists between thedifferential signals SSI and/SSI.

In FIGS. 29A to 29C, the horizontal axis indicates time, and thevertical axis indicates voltage. FIG. 29A indicates the eye patterns bythe signal voltage waveforms at the input terminal of the receptionbuffer circuit 1003P. In FIG. 29A, the dotted waveform SSIB indicatesthe signal voltage waveform generated by the components of thedifferential signals SSI and/SSI. In FIG. 29A, the solid-line waveformSKYB is a signal voltage waveform generated by the skew.

FIG. 29B is a diagram illustrating eye patterns formed by the signalvoltage waveform SSIB extracted from FIG. 29A. That is, FIG. 29B is adiagram illustrating the eye patterns formed by the signal voltagewaveform SSIB generated by the components of the differential signalsSSI and/SSI. FIG. 29C is a diagram illustrating the eye patterns formedby the signal voltage waveform SKYB extracted from FIG. 29A. That is,FIG. 29C is a diagram illustrating an eye pattern formed by the signalvoltage waveform CMIB generated by the skew.

The eye patterns formed by the signal voltage waveform SSIB are in anopen state as illustrated in FIGS. 29A and 29B and, on the other hand,it is difficult to recognize an open part in the eye pattern formed bythe signal voltage waveform CMIB. The components of the differentialsignal are equalized by the digital filter 1002P, and the deformedwaveform is shaped. By the shaping, the eye pattern formed by the signalvoltage waveform SSIB becomes an open state as illustrated in FIG. 29B,and the logic values of the differential signals expressed by thedifferential signals SSI and/SSI can be identified and specified withhigh precision.

On the other hand, the components corresponding to the common modesignal in the skew are transmitted to the reception buffer circuit 1003Pwithout being equalized by the digital filter 1002P. Consequently,although the external shape of the eye pattern of the signal voltagewaveform SKYB supplied to the input terminal of the reception buffercircuit 1003P can be determined as illustrated in FIG. 29C, it isdifficult to recognize an open eye pattern. It can prevent the signalvoltage waveform generated by the skew from being erroneously identifiedas the logic value of the differential signal.

Delay Element

Next, the delay element DLN as a component of the digital filter 1002Pwill be described. The delay element DLN is comprised of a delay linecoupled between the ends SNO1 and SNO2 of the signal lines 1000P1 and1000P2 near the input terminal of the reception buffer circuit. Thesignal loss per unit length of the delay line is higher than that of thesignal lines 1000P1 and 1000P2.

As a method of increasing the signal loss, for example, it is regardedthat the delay line as a component of the delay element DLN is comprisedof a plurality of distributed constant circuits as illustrated in FIG.27B and the value of a series resistance R in the plurality ofdistributed constant circuits corresponding to the unit length of thedelay line is made higher than that of the signal line and/or, in aplurality of distributed constant circuits corresponding to the unitlength, parallel resistance for the predetermined voltage Vs is madelower than that of the signal line. It corresponds to, for example, inFIG. 27B, increase in the conductance G in the distributed constantcircuit.

The round-trip signal delay in the delay element DLN comprised of thedelay line is desirably set to a value obtained by dividing the datawidth interval UT by the coefficient mm. That is, the round-trip signaldelay is desirably set to UT/mm. The coefficient mm is ½ or an integer“m”. The integer m is, for example, a natural number of 1, 2, 3, or thelike.

As described above, one differential signal SSI (or the otherdifferential signal/SSI) of the differential signals SSI and/SSI can beregarded as the reference potential. In this case, the equivalentcircuit of the delay element illustrated in FIG. 27B becomes equal tothe equivalent circuit (FIG. 1B) described in the first embodiment whenthe differential signals are supplied to the ends DNIO1 and DNIO2.Consequently, the equalizing function works on the components of thedifferential signals excluding the common mode signals and the like inthe differential signals SSI and/SSI as described in the firstembodiment, so that the deformed waveform can be restored.

To equalize the equivalent circuit illustrated in FIG. 27B and thatillustrated in FIG. 1B, for example, when the reference potential (thepredetermined voltage Vs in the first embodiment) is supplied to the endDNIO2 in the equivalent circuit illustrated in FIG. 27B, it is necessaryto set a complex impedance viewed from the end DNIO1 using the end DNIO2as a reference as the equation (14) in FIG. 30. Z_(DF) indicates acomplex impedance of the delay element DLN illustrated in FIG. 27A, andZ_(SE) indicates a complex impedance of the delay element DLNillustrated in FIG. 1A.

In the equation (14), the right side indicates that two delay elementshaving the same complex impedance Z_(SE) are coupled in series.Consequently, to make the equivalent circuit of FIG. 27B and that ofFIG. 1B equal, each of the loss and delay of the delay element DLNillustrated in FIG. 27A is twice as large as that of the delay elementDLN illustrated in FIG. 1A. In the seventh embodiment, consequently,different from the first embodiment, the round-trip signal delay in thedelay element DLN illustrated in FIG. 27A starts from the half (½) ofthe data width interval UT, not one. That is, the coefficient mmdetermining the round-trip signal delay is an integer starting from ½ or1, not an integer starting from 1.

In the seventh embodiment, when the common mode signal (in-phase signal)is transmitted to the pair of signal lines (differential signal lines)1000P1 and 1000P2, since both ends of the delay element DLN have equalpotentials, there is no signal transmitted to the delay element DLN.Therefore, the delay element DLN does not perform digital filteroperation performing equalization. On the other hand, when differentialsignals are transmitted, a potential difference occurs between both endsof the delay element DLN, so that the delay element DLN performs thedigital filter operation performing equalization in a manner similar tothe first embodiment and the like.

That is, in the seventh embodiment, the delay element DLN is a digitalfilter selectively working only on the components of the differentialsignals.

Further, in the seventh embodiment, equalizer performance can beimproved. That is, since serial communication is performed by thedifferential signals, direct-current coupling can be eliminated.Consequently, the influence of noise in the power source voltage and theground voltage (predetermined voltage Vs) can be avoided, and theequalizer performance can be improved. In addition, so-called limitperformance can be also improved.

In the seventh embodiment, in reality, limit time is necessary forreflection of the differential signals and combination of thedifferential signals at each of ends of the delay element DLN. There is,so-called frequency dispersion that the time slightly varies dependingon frequency at which reflection/combination occurs. The frequency bandnecessary for reception, of a digital signal is expressed by theequation (15) in FIG. 30. N denotes maximum data length. When thefluctuation range of time required to reflection and combination of asignal in the frequency band is Δt and signal delay of the delay elementDNL is τ, even if the frequency dispersion of the delay amount is zero,at the time of reflection of a signal, a relative timing error expressedby the equation (16) in FIG. 30 occurs. In the case of usingdifferential signals as described in the seventh embodiment, the signaldelay τ becomes twice, so that the relative timing error becomes thehalf. The equation (16) indicates limit performance. When a filter isclose to ideal, the relative timing error of the digital filterdescribed in the seventh embodiment is reduced as compared with that ofthe digital filter described in the first embodiment.

Structure of Digital Filter

Next, the structure of the digital filter 1002P according to the seventhembodiment will be described. FIG. 31 is a plan view when the part ofthe broken-line region DFA is seen from the first main surface PPF1 ofthe interposer PPS-1 in FIG. 6. FIG. 32 is the A5-A5′ section in FIG.31. In FIGS. 31 and 32, the delay element DLN as a component of thedigital filter 1002P is comprised of the conductive layer formed in theinterposer PPS-1.

FIGS. 31 and 32 are similar to FIGS. 7 and 8 described above. The pointsdifferent from FIGS. 7 and 8 will be mainly described. First, in FIGS. 7and 8, the case of the signal end that the serial signal propagatingthrough the signal line changes using the predetermined voltage Vs as areference potential has been described. On the other hand, in theseventh embodiment, differential signals are transmitted by using a pairof signal lines. Accordingly, each of the transmission buffer circuitsSCB1-1 and SCB2-1 and the reception buffer circuits RCB1-1, RCB1-2,RCB2-1, and RCB2-2 illustrated in FIG. 6 is comprised of a differentialcircuit.

In FIG. 31, SSN-L30 and SSP-L30 indicate a pair of signal lines (wiringpatterns) transmitting a pair of differential signals, and SSG-L30indicates a voltage wire (wiring pattern) to which the predeterminedvoltage Vs is supplied. In FIG. 31, SSD-L10 indicates a delay line(wiring pattern) as a component of the delay element DLN, and SSG-L10indicates a voltage wire (wiring pattern) to which the predeterminedvoltage Vs is supplied.

In the seventh embodiment, the delay line SSD-L10 is formed by the firstconductive layer INS-L1 formed in the interposer INS-1 which will bedescribed later with reference to FIG. 32. The voltage wire SSG-L10 isalso formed by the first conductive layer INS-L1 formed in theinterposer INS-1. That is, both of the delay line SSD-L10 and thevoltage wire SSG-L10 are formed by the same first conductive layer. Inplan view, the voltage wire SSG-L10 is close to the delay line SSD-L10and extends in parallel to the delay line SSD-L10. Since the voltagewire SSG-L10 and the delay line SSD-L10 are disposed so as to be apartfrom each other, the voltage wire SSG-L10 and the delay line SSD-L10 areelectrically separated. Between the delay line SSD-L10 and the voltagewire SSG-L10 disposed closely, the capacitance C and the conductance Gillustrated in FIG. 27B are formed.

One end of the delay line SSD-L10 is coupled to the signal line SSP-L30via the contact CT2, and the other end is coupled to the signal lineSSN-L30 via the contact CT2. The signal line SSN-L30 is coupled to themicrobump electrode INS-MPD2N via the contact CT2, and the signal lineSSP-L30 is coupled to the microbump electrode INS-MPD2P via the contactCT2. Each of the microbump electrodes INS-MPD2N and INS-MPD2P is coupledto a corresponding microbump electrode formed over a second main surfaceCLF2 of a not-illustrated logic semiconductor chip LCH-1 by themicrobumps MBM-SN and MBM-SP and coupled to a pair of input terminals ofthe reception buffer circuit RCB1-2 formed in the semiconductor regionSS.

To the signal lines SSN-L30 and SSP-L30, a pair of differential signalsis supplied from the logic semiconductor chip LCH-2 via anot-illustrated print substrate PBS. Consequently, a pair ofdifferential signals from the logic semiconductor chip LCH-2 propagatesthrough the pair of signal lines SSN-L30 and SSP-L30 and is supplied tothe pair of input terminals of the reception buffer circuit RCB1-2 ofthe logic semiconductor chip LCH-1.

In plan view, one end of the voltage wire SSG-L10 disposed along thesignal line SSD-L10 is coupled to the voltage wire SSG-L30 via thecontact CT2 and the other end is coupled to the voltage wire SSG-L30 viathe contact CT so as to surround the delay line SSD-L10. The voltagewire SSG-L30 is coupled to the corresponding microbump electrodeINS-MPD1 via the contact CT2, and each of the microbump electrodesINS-MPD1 is coupled to the corresponding microbump electrode formed overthe second main surface CLF2 of the not-illustrated logic semiconductorchip LCH-1 by the microbump MBM-G.

To the voltage wire SSG-L30, the predetermined voltage Vs is suppliedfrom the print substrate PBS via a wire formed in a not-illustratedpackage substrate PPS-1. Consequently, the predetermined voltage Vs issupplied to the logic semiconductor chip LCH-1 and also to the voltagewire SSG-L10.

FIG. 32 is the A5-A5′ section in FIG. 31. In FIG. 32, the printsubstrate PBS, the package substrate PPS-1, and the like are the same asthose in FIG. 8, the description will not be repeated. A pair ofdifferential signals from the logic semiconductor chip LCH-2 aretransmitted to the bump electrode PPS-MPD formed over the first mainsurface PPF1 of the package substrate PPS-1 via the print substrate PBSand the package substrate PPS-1. The bump electrode PPS-MPD is coupledto the bump electrode INS-SPD formed over the second main surface INF2of the interposer INS-1 by the bump electrode SMB.

The bump electrode INS-SPD is coupled to the wiring pattern INS-L1(R)comprised of the first conductive layer INS-L1 formed in the interposerINS-1 via the contact CT2S formed over the silicon substrate SSB-1, andthe wiring pattern INS-L1(R) is coupled to the wiring pattern INS-L2(R)formed by the second conductive layer INS-L2 via the contact CT2. Thewiring pattern INS-L2(R) is coupled to the signal line SSP-L30 as awiring pattern formed by the third conductive layer INS-L3 via thecontact CT2.

The signal line SSP-L30 extends in the lateral direction in FIG. 32. Inthe extended part, the signal line SSP-L30 is coupled to the microbumpelectrode INS-MPD2P via the contacts CT2. The signal line SSP-L30 iscoupled to the wiring pattern INS-L2(R) formed by the second conductivelayer INS-L2 via the contacts CT2, and the wiring pattern INS-L2(R) iscoupled to one end of a wiring pattern corresponding to the delay lineSSD-L10 via the contact CT2. The delay line SSD-L10 is comprised of thefirst conductive layer INS-L1 formed in the interposer INS-1.

The microbump electrode INS-MPD2P is coupled to the microbump electrodeLCH-PD3 formed on the first main surface CHF2 of the logic semiconductorchip via the microbump MBM-SP, and the microbump electrode LCH-PD3 iscoupled to the semiconductor region SS via the wiring patterns LCH-L1(R)to LCH-L3(R) in the wiring layer HSB formed on the main surface of thesilicon substrate SSB and the contacts CT1. In the semiconductor regionSS, the reception buffer circuit RCB102 is formed. Differential signalsfrom the logic semiconductor chip LCH-2 are supplied to the inputterminal of the reception buffer circuit RCB1-2.

In the seventh embodiment, the round-trip signal delay UT/mm isdetermined by delay time of a signal transmitted between one end and theother end of the delay line SSD-L10. To obtain desired round-trip signaldelay, for example, the length between one end and the other end of thedelay line SSD-L10 and the line width of the delay line SSD-L10 are set.As illustrated in FIG. 31, also in the seventh embodiment, the linewidth BLD1 of the delay line SSD-L10 is narrower than the line width BLSof the signal lines SSP-L30 and SSN-L30. When it is regarded that thedelay line SSD-L10 and the signal lines SSP-L30 and SSN-L30 areintegrated, the line having the narrow width using the part when theline width changes corresponds to the delay line SSD-L10, and the widelines sandwiching the narrow line are the signal lines SSP-L30 andSSN-L30.

Although only the section related to the coupling part of the signalline SSP-L30 and one end of the delay line SSD-L10 is illustrated inFIG. 32, the coupling between the signal line SSN-L30 and the other endof the delay line SSD-L10 is similar. In the seventh embodiment, thevoltage wire SSG-L10 is formed by the same conductive layer as that ofthe delay line SSD-L10. Consequently, the coupling part between thevoltage wires SSG-L30 and SSG-L10 is similar to that in FIG. 32.

In the seventh embodiment, the delay line as a component of the delayelement DLN is formed by the first conductive layer INS-L1 coupled to apair of signal lines SSN-L30 and SSP-L30 near the pair of microbumpelectrodes INS-MPD2N and INS-MPD2P to which the pair of signal linesSSN-L30 and SSP-L30 transmitting a pair of differential signals arecoupled via the contacts CT2 and the like. Since each of the pair ofsignal lines SSN-L30 and SSP-L30 is formed by the third conductive layerINS-L3, the delay line SSD-L10 and the signal lines SSN-L30 and SSP-L30are formed by different conductive layers, and the delay line SSD-L10 iscoupled between the pair of microbump electrodes INS-MPD2N andINS-MPD2P.

A signal propagating through the delay line SSD-L10 is a single-endsignal which changes relative to the reference potential. If the voltagewire supplying the reference potential is not specified, when the delayline SSD-L10 is expressed by an equivalent circuit as illustrated inFIG. 27B, it becomes difficult to specify, for example, the values ofthe capacitance C and the conductance G. Consequently, in the seventhembodiment, the voltage wire SSG-L10 to which the predetermined voltageVs as the reference potential is supplied is formed by the firstconductive layer INS-L1 which is the same as that of the delay lineSSD-L10, and the voltage line SSG-L10 is disposed in parallel to thedelay line SSD-L10 near the delay line SSD-L10. In plan view, althoughnot limited, the line width BLD2 of the voltage wire SSG-L10 is set tothe same as the line width BLD1 of the delay line SSD-L10 and is set tobe narrower than the line width BLS of each of the signal lines SSN-L30and SSP-L30 and the voltage wire SSG-L30. When the data width intervalUT is set to one data unit of a serial signal, the round-trip signaldelay of the delay element DLN becomes UT/mm. mm denotes ½ or a naturalnumber (integer starting from 1). When mm is set to ½, the round-tripsignal delay corresponds twice as large as the data width interval UT.

In FIG. 31, the signal lines SSN-L30 and SSP-L30 and a part of the delayline SSD-L10 are disposed so as to be linear, and the signal linesSSN-L30 and SSP-L30 and a part of the voltage wire SSG-L10 are disposedso as to be linear. Since the conductive layer forming the signal linesSSN-L30 and SSP-L30 is different from the conductive layer forming thedelay line SSD-L10 and the voltage wire SSG-L10, the lines may not belinear. That is the disposition relations between the signal linesSSN-L30 and SSP-L30 and the delay line SSD-L10 and the voltage wireSSG-L10 are arbitrary. For example, the delay line SSD-L10 and thevoltage wire SSG-L10 may be disposed so as to be orthogonal to thesignal lines SSN-L30 and SSP-L30.

Modification

FIGS. 33 and 34 are a plan view and a cross section illustrating thestructure of a digital filter according to a modification of the seventhembodiment. FIG. 34 is a cross section A6-A6′ in FIG. 33. FIGS. 33 and34 are similar to FIGS. 31 and 32. The different points will be mainlydescribed.

In FIGS. 31 and 32, each of the delay line SSD-L10 as a component of thedelay element DLN and the voltage wire SSG-L10 disposed along the delayline SSD-L10 is formed by the first conductive layer INS-L1 differentfrom the signal lines SSN-L30 and SSP-L30 and the voltage wire SSG-L30.In contrast, in FIGS. 33 and 34, each of the delay line SSD-L10 and thevoltage wire SSG-L10 is formed by the third conductive layer INS-L3which is the same as that of the signal lines SSN-L30 and SSP-L30 andthe voltage wire SSG-L30. In FIGS. 33 and 34, the delay line isexpressed as SSD-L30, and the voltage wire disposed along the delay lineSSD-L30 is expressed as SSG-L32.

Since the signal lines SSN-L30 and SSP-L30 and the delay line SSD-L30are formed by the same third conductive layer INS-L3, those lines areintegrated. In FIGS. 31 and 32, to couple the delay line SSD-L10 and thesignal lines SSN-L30 and SSP-L30, the contacts CT2 and the wiringpattern INS-L2(R) are necessary. In FIGS. 33 and 34, the contacts CT2and the wiring pattern INS-L2(R) are not provided.

Similarly, since the voltage wires SSG-L30 and SSG-L32 are formed by thesame third conductive layer INS-L3, those wires are integrated. In FIGS.31 and 32, to couple the voltage wires SSG-L30 and SSG-L10, the contactsCT2 and not-illustrated wiring pattern are necessary. In FIGS. 33 and34, however, the contacts CT2 and the wiring pattern are not provided.

Although the signal lines SSN-L30 and SSP-L30 and the delay line SSD-L30are integrated, the border between the signal lines and the delay linecan be specified as the part where the line width of the lines changes.In this case, the part in which the line width of the line changes fromBLS to BLD1 and the part in which the line width changes from BLD1 toBLS are the border between the signal lines and the delay line. Theregion of the line whose width is narrow like BLD1 corresponds to thedelay line SSD-L30. At this time, the regions in which the line width ofthe line is BLS correspond to the signal lines SSN-L30 and SSP-L30.

Similarly, the border between the voltage wires SSG-L30 and SSG-L32 canbe specified as the part in which the line width of the wire changes. Inthis case, the region in which the line width of the line is narrowerthan the line width BLS and is BLD2 corresponds to the voltage wireSSG-L32 disposed along the delay line SSD-L30, and the region in whichthe line width of the wire is BLS corresponds to the voltage wireSSG-L30.

Since the signal line SSP-L30 and the delay line SSD-L30 are integrated,in FIG. 34, the wiring patter formed by the third conductive layerINS-L3 extends in the lateral direction and becomes the signal wireSSP-L30 and the delay line SSD-L30. In FIG. 34, only the signal lineSSP-L30 and the delay line SSD-L30 integrally formed by the wiringpattern are illustrated. The other signal lines and voltage wires arealso similar to those of FIG. 34. That is, the signal line SSN-L30 andthe delay line SSD-L30 are also integrally formed by the thirdconductive layer INS-L3. The voltage wires SSG-L30 and SSG-L32 are alsointegrally formed by the third conductive layer INS-L3.

In the modification, the delay line SSD-L30 and the voltage wire SSG-L32are formed by the same conductive layer as that of the signal linesSSN-L30 and SSP-L30 and the voltage wire SSG-L30. Consequently, thenumber of conductive layers configuring the digital filter 1002P can bereduced. In other words, the conductive layers configuring the digitalfilter 1002P can be saved.

Although not limited, in the seventh embodiment, the microbump MBM-SP ismade of copper (Cu), and the bump SBL is a solder ball.

The correspondence between the structure of the digital filter describedwith reference to FIGS. 31 to 34 and the digital filter illustrated inFIGS. 1 and 27 will be described as follows.

The nodes WRN1 and WRN2 as the wired-OR coupling parts illustrated inFIG. 27 corresponds to the contacts CT2 coupling the signal linesSSN-L30 and SSP-L30 and the delay line SSD-L10 in FIGS. 31 and 32. InFIGS. 33 and 34, the borders between the signal lines SSN-L30 andSSP-L30 and the delay line SSD-L30 correspond to the nodes WRN1 and WRN2illustrated in FIG. 27. That is, the border regions in which the linewidth of the line changes correspond to the nodes WRN1 and WRN2.

When only the components of a pair of differential signals propagatethrough the pair of signal lines SSN-L30 and SSP-L30 and reach the nodesWRN1 and WRN2 as the wired coupling parts, reflection of the signal isrepeated between the nodes WRN1 and WRN2. Specifically, multiple signalreflection occurs and the signals are combined at the nodes WRN1 andWRN2. Consequently, the digital filter 1002P equivalently calculates theequation (1) in FIG. 1C with respect to the components of thedifferential signals. It means that, for example, an inverse transferfunction is equivalently calculated with respect to the transferfunction of the signal line coupling the transmission buffer circuit andthe reception buffer circuit provided for different logic semiconductordevices. Since the digital filter 1002P does not have an active element,only attenuation occurs in equivalent calculation of the inversetransfer function. Consequently, calculation of an inverse transferfunction deviated only by the amount of a constant corresponding to theattenuation is executed as equivalent calculation of an inverse transferfunction.

In the seventh embodiment, with respect to the components of thedifferential signals, the above-described coefficient mm is used inplace of the coefficient “m” in the equation (1). That is, not only aninteger but also ½ can be substituted as the coefficient “m” for theequation (1).

In the seventh embodiment, the voltage wire SSG-L10 (SSG-L32) formed inthe same layer as that of the delay line SSD-L10 (SSD-L30) is disposedclose to the delay line SSD-L10 (SSD-L30). A signal loss per unit lengthof each of the delay line SSD-L10 (SSD-L30) and the voltage wire SSG-L10(SSG-L32) is set to be larger than the signal loss per unit length ofthe signal lines SSN-L30 and SSP-L30. For example, the boundary lengthin section of each of the delay line SSD-L10 (SSD-L30) and the voltagewire SSG-L10 (SSG-L32) is set to be smaller than that of the signallines SSN-L30 and SSP-L30. By adjusting the signal losses in thoselines, for example, the inductance L and the resistance R in thedistributed constant circuits illustrated in FIGS. 1C and 27B can beadjusted to arbitrary values. By adjusting the interval between thedelay line SSD-L10 (SSD-L30) and the voltage wire SSG-L10 (SSG-L32), thecapacitance C and the conductance G in the distributed constant circuitcan be adjusted to arbitrary values. Obviously, the inductance L, theresistance R, the capacitance C, and the conductance G can be adjustedby adjusting the boundary length in section and the interval (includingthe line width) of the delay line SSD-L10 (SSD-L30) and the voltage wireSSG-L10 (SSG-L32). In such a manner, an arbitrary inverse transferfunction can be equivalently calculated by the delay line.

Eighth Embodiment

FIGS. 35 and 36 are a plan view and a cross section illustrating thestructure of a digital filter according to an eighth embodiment. FIG. 36is an A7-A7′ section in FIG. 35. FIGS. 35 and 36 are similar to FIGS. 31to 34. The points different from FIGS. 31 and 32 will be mainlydescribed.

In FIG. 35, SSD-L10 indicates a delay line as a component of the digitalfilter 1002P, and SSG-L32 indicates a voltage wire to which thepredetermined voltage Vs is supplied. In the eighth embodiment, asillustrated in FIG. 36, the delay line SSD-L10 is formed by the firstconductive layer in the three conductive layers formed in the interposerINS-1. The voltage wire SSG-L32 is formed by the third conductive layer.The voltage wire SSG-L32 is disposed so that its part overlaps in thedelay line SSD-L10 in plan view as illustrated in FIG. 35. Specifically,in the seventh embodiment, in plan view, the voltage wire SSG-L10(SSG-L32) is disposed close to and parallel to the delay line SSD-L10(SSD-L30). On the other hand, in the eighth embodiment, the voltage wireSSG-L32 is disposed stereoscopically close to and parallel to the delayline SSD-L10. Obviously, an insulating layer exists between the voltagewire SSG-L32 and the delay line SD-L10, and the voltage wire SSG-L32 andthe delay line SD-L10 are electrically separated.

Also in the eighth embodiment, the ends of the delay line SSD-L10 arecoupled to the pair of signal lines SSN-L30 and SSP-L30 via the contactsCT2. The voltage wire SSG-L32 is formed integrally with the voltage wireSSG-L30.

Also in the eighth embodiment, the delay line SSD-L10 and the voltagewire SSG-L32 are set so that the signal loss per unit length becomeslarger than that of the pair of signal lines SSN-L30 and SSP-L30. Forexample, the boundary length in section of the delay line SSD-L10 andthe voltage wire SSG-L32 is set to be smaller than that of the signallines SSN-L30 and SSP-L30. In the example of FIG. 35, the boundarylength in section is decreased by making the line width of the delayline SSD-L10 narrower than that of the signal lines SSN-L30 and SSP-L30.The boundary length in section of the voltage wire SSG-L32 is decreasedby making the thickness of the voltage wire SSG-L32 smaller than that ofthe signal line SSD-L10.

In the eighth embodiment, for example, by adjusting the boundary lengthin section of the delay line SSD-L10 and the voltage wire SSG-L32 andthe distance (interlayer distance) between the overlapped regions, theinductance L, the resistance R, the capacitance C, and the conductance Gin the distributed constant circuit are adjusted. Since the operation ofthe digital filter 1002P according to the eighth embodiment is similarto that of the seventh embodiment, the description will not be repeated.

The line width of the delay line SSD-L10 becomes narrower than that ofthe signal lines SSN-L30 and SSP-L30 and the like. The voltage lineSSG-L32 overlaps the delay line SSD-L10. Consequently, in plan view, thedigital filter 1002P can be disposed in a small area. Therefore, theeighth embodiment is suitable in the case where, for example, in planview, the interposer INS-1 does not have much area. On the other hand,the digital filter 1002P described in the seventh embodiment is suitableto the case where the number of conductive layers formed in theinterposer INS-1 is limited.

Ninth Embodiment

FIGS. 37 and 38 are a plan view and a cross section illustrating thestructure of a digital filter according to a ninth embodiment. In theninth embodiment, a structure that the logic semiconductor chip LCH-1,not the interposer INS-1, has the digital filter 1002P is provided. FIG.37 is a plan view when the logic semiconductor device LCH-1 is seen fromthe second main surface side CHF2. FIG. 38 is an A8-A8′ cross section inFIG. 37. In FIG. 38, the section of the package substrate PPS-1 and theprint substrate PBS in the A8-A8′ cross section is also illustrated.

In the ninth embodiment, in a manner similar to the third embodiment,the wiring pattern as a component of the delay element DLN is formed inthe logic semiconductor chip LCH-1. Since the delay element DLN as acomponent of the digital filter 1002P is formed in the logicsemiconductor chip LCH-1, although the example of the semiconductordevice which does not use the interposer INS-1 will be escribed,obviously, the interposer INS-1 may be provided between the logicsemiconductor chip LCH-1 and the package substrate.

In FIG. 38, PPS-1 indicates the package substrate, and PBS denotes theprint substrate. Since the structure of the package substrate PPS-1 andthe print substrate PBS has been described, for example, in FIG. 15related to the third embodiment, the description will not be repeated.The logic semiconductor chip LCH-1 is mounted over the package substratePPS-1 so that its second main surface CHF2 faces the first main surfacePPF1 of the package substrate PPS-1. The bump electrode PPS-MPD formedon the first main surface PPF1 of the package substrate PPS-1 is coupledto a microbump electrode formed on the second main surface CHF2 of thelogic semiconductor chip LCH-1. In the diagram, the microbump electrodeis indicated as LCH-PDP.

The logic semiconductor chip LCH-1 has the silicon substrate SSB inwhich a semiconductor region for configuring an element and the like isformed, and the wiring layer HSB formed on a main surface of the siliconsubstrate SSB. The wiring layer HSB has a plurality of conductive layersand a plurality of insulating layers alternately stacked. In the ninthembodiment, the wiring layer HSB has three conductive layers (wiringlayers). In FIG. 38, LCH-L10(R) is a wiring pattern formed by the firstconductive layer, LCH-L20(R) is a wiring pattern formed by the secondconductive layer, and LCH-L30(R) is a wiring pattern formed by the thirdconductive layer. CT1 indicates contacts electrically coupling theconductive layers via the insulating layers provided between theconductive layers.

In FIG. 38, WEL indicates a well region formed in the silicon substrateSSB. The well region WEL has a conduction type opposite to that of thesilicon substrate SSB. For example, when the silicon substrate SSB isN-type semiconductor, the well region WEL is a P-type semiconductorregion. In the diagram, GIO indicates an insulating film. The insulatingfilm GIO is formed on the main surface of the well region WEL, and thewiring pattern LCH-L10(R) is formed on the insulating film GIO. Theinsulating film GIO is, for example, a gate insulating film of MOSFET,and the wiring pattern LCH-L10(R) is a gate electrode formed on the gateinsulating film.

FIG. 37 is a plan view from the direction (visual direction) indicatedby the arrow A8 in FIG. 38 and, that is, a plan view when the logicsemiconductor chip LCH-1 is seen from the second main surface CHF2. InFIG. 37, LCH-PDG indicates microbump electrodes which are coupled to thebump electrodes PPS-MPD formed on the package substrate PPS-1 via thebump SMB and to which the predetermined voltage Vs is supplied via thepackage substrate PPS-1.

In FIG. 37, LCH-PDN and LCH-PDP indicate microbump electrodes which arecoupled to the bump electrode PPS-MPD formed on the package substratePPS-1 via the sump SMB and to which a pair of differential signals issupplied from the print substrate PBS via the package substrate PPS-1.That is, a pair of differential signals is supplied to the microbumpelectrodes LCH-PDN and LCH-PDP using the conductive layers formed in theprint substrate PBS and the package substrate PPS-1 as a pair of signalpaths.

The microbump electrode LCH-PDG is coupled to the voltage wire SSG-L30comprised of the third conductive layer formed in the wiring layer HSBof the logic semiconductor chip LCH-1. The voltage wire SSG-L30 isohmic-coupled to the well region WEL via the contact CT1. With theconfiguration, the predetermined voltage Vs is supplied to anot-illustrated circuit block in the logic semiconductor chip LCH-1 viathe voltage wire SSG-L30 and, the predetermined voltage Vs is suppliedalso to the well region WEL.

The microbump electrode LCH-PDN is coupled to the signal line SSN-L30formed by the third conductive layer formed in the wiring layer HSB inthe logic semiconductor chip LCH-1 via the contact CT1. The microbumpelectrode LCH-PDP is coupled to the signal line SSP-L30 formed by thethird conductive layer formed in the wiring layer HSB of the logicsemiconductor chip LCH-1 via the contact CT1. The signal lines SSN-L30and SSP-L30 are coupled to a pair of input terminals of a receptionbuffer circuit (corresponding to the reception buffer circuit describedin the seventh embodiment) provided in the logic semiconductor chipLCH-1. With the configuration, a pair of differential signals fromanother logic semiconductor chip (for example, LCH-2 illustrated in FIG.6) is supplied to a pair of input terminals of a reception buffercircuit via signal lines formed by conductive layers in the printsubstrate PBS, the package substrate PPS-1, and the like.

Each of the signal lines SSN-L30 and SSP-L30 is coupled to the delayelement DLN as a component of the digital filter 1002P. In the ninthembodiment, the delay element DLN has the delay line SSD-L10 formed bythe first conductive layer in the conductive layer HSB of the logicsemiconductor chip LCH-1, one end of the delay line SSD-L10 is coupledto the signal line SSP-L30, and the other end of the delay line SSD-L10is coupled to the signal line SSN-L30.

Coupling between the signal line SSP-L30 and the delay line SSD-L10 willbe described with reference to FIG. 38. In FIG. 38, the wiring patternLCH-L30(R) formed by the third conductive layer corresponds to thesignal line SSP-L30 illustrated in FIG. 37. In FIG. 38, the wiringpattern LCH-L10 formed by the first conductive layer corresponds to thedelay line SSD-L10. The signal line SSP-L30 (LCH-L30(R)) is coupled tothe wiring pattern LCH-L20(R) formed by the second conductive layer viathe contact CT1, and the wiring pattern LCH-L20(R) is coupled to one endof the delay line SSD-L10 (LCH-L10(R)) via the contact CT1. In FIG. 37,to avoid complication of the drawing, the wiring pattern LCH-L20(R) isnot drawn.

Similarly, the signal line SSN-L30 is coupled to the other end of thedelay line SSD-L10 (LCH-L10(R)). With the configuration, the delay lineSSD-L10 disposed over the well region WEL to which the predeterminedvoltage Vs is supplied is coupled between the pair of signal linesSSN-L30 and SSP-L30 to which a pair of differential signals is suppliedvia the insulating film GIO.

In the ninth embodiment, the insulting film GIO is interposed betweenthe wiring pattern LCH-L10(R) corresponding to the delay line SSD-L10and the well region WEL to which the predetermined voltage Vs issupplied. Therefore, a MOS capacitive element using the insulating filmGIO as a dielectric and using the delay line SSD-L10 and the well regionWEL as electrodes is formed. The MOS capacitive element can beequivalently regarded as a MOS diode element. Consequently, when thedelay line SSD-L10 is regarded as the equivalent circuit illustrated inFIG. 27B, by adjusting not only the resistance R of the wiring patternLCH-L10(R) but also the capacitance C and the conductance G equivalentlyformed between the wiring pattern LCH-L10(R) and the well region WEL,the signal loss amount in the delay line SSD-L10 can be controlled. Theinsulating film GIO is formed by, for example, the gate insulating filmof the MOSFET. In this case, since the dielectric constant of thesilicon substrate is high, the delay amount per unit length of the delayline SSD-L10 can be increased, and the digital filter 1002P can beminiaturized.

Modification

FIGS. 39 and 40 are a plan view and a cross section illustrating thestructure of a digital filter 1002P according to a modification of theninth embodiment. Like FIG. 37, FIG. 39 is a plan view when the logicsemiconductor chip LCH-1 is seen from the second main surface CHF2 side.The visual direction is expressed as the arrow A9 in FIG. 40. FIG. 40 isan A9-A9′ cross section in FIG. 39. Like FIG. 38, FIG. 40 alsoillustrates the section of the package substrate PPS-1 and the printsubstrate PBS in the A9-A9′ section.

Since FIGS. 39 and 40 are similar to FIGS. 37 and 38, the differentpoints will be mainly described. In FIGS. 37 and 38, the predeterminedvoltage Vs is supplied to the well region WEL formed in the siliconsubstrate SSB, and the wiring pattern LCH-L10(R) disposed above the wellregion WEL via the insulating film GIO is used as the delay lineSSD-L10.

In the modification illustrated in FIGS. 39 and 40, in the well regionWEL to which the predetermined voltage Vs is supplied, a semiconductorregion of a conduction type opposite to that of the well region WEL isformed. To the semiconductor region, the wiring pattern LCH-L10(R) isohmic-coupled. The wiring pattern LCH-L10(R) is used as the delay lineSSD-L10 as a component of the delay element DLN.

In FIG. 40, DFR indicates a semiconductor region formed in the wellregion WEL. For example, when the well region WEL is a P-typesemiconductor region, the semiconductor region DFR is an N-typesemiconductor diffusion region formed in the well region WEL. In planview, the semiconductor region DFR is formed in a U shape as illustratedin FIG. 37. By the first conductive layer in the wiring layer HSB, theU-shaped wiring pattern LCH-L10(R) which overlaps the semiconductorregion DFR is formed (in FIG. 39, indicated as delay line SSD-L10). Thewiring pattern LCH-L10(R) is ohmic-coupled to the overlappedsemiconductor region DFR (FIG. 40).

As illustrated in FIG. 40, one end of the wiring pattern LCH-L10(R) iscoupled to the wiring pattern LCH-L30(R) as the signal line SSP-L30 viathe contact CT1 and the wiring pattern LCH-L20(R) in the wiring layerHSB in the second layer. Similarly, the other end of the wiring patternLCH-L10(R) is coupled to the wiring pattern as the signal line SSN-L30via the contact CT1 and the wiring pattern in the second layer. In FIG.39, to avoid complication of the diagram, the wiring pattern (forexample, the wiring pattern LCH-L20(R)) formed by the second conductivelayer is not drawn.

In the modification, since the semiconductor region DFR is formed in thewell region WEL, a PN-junction diode element is formed by thesemiconductor region DFR and the well region WEL. Since thepredetermined voltage Vs is supplied to the well region WEL and thedelay line SSD-L10(LCH-L10(R)) is ohmic-coupled to the semiconductorregion DFR, the PN-junction diode element is coupled between thepredetermined voltage Vs and the delay line SSD-L10. In the case ofseeing the equivalent circuit (FIG. 27) of the delay element DLN,junction current flows in the PN junction diode. Consequently, theconductance G can be increased. Therefore, the signal loss in the delayelement DLN can be increased, and the digital filter 1002P can beminiaturized. By controlling the inverse bias voltage supplied to the PNjunction diode, the conductance G can be controlled, and the signal lossin the delay element DLN can be controlled.

Although relative permittivity of an oxide film is about 4, relativepermittivity of silicon becomes about 12. Consequently, delay in thedelay element DLN can be increased to about 1.7 times as the positivesquare root of (12/4), and the digital filter 1002P can be miniaturized.

In the ninth embodiment and its modification, when the resistance of thewiring pattern LCH-L10(R) is higher than a desired resistance value, forexample, as illustrated in FIG. 16C, it is sufficient to couple a wiringpattern formed by the second conductive layer to the wiring patternLCH-L10(R) in parallel.

In the ninth embodiment, the resistance R per unit length of the delayline SSD-L10 relative to the predetermined voltage Vs is made smallerthan that of the signal line, and the conductance G per unit length ofthe delay line SSD-L10 to the predetermined voltage Vs is made higher.

Although the example of forming the digital filter in the logicsemiconductor chip LCH-1 has been described, the invention is notlimited to the example. For example, in the case of using a siliconinterposer as the interposer INS-1, the well region WEL, thesemiconductor region DFR, and the wiring pattern LCH-L10(R) describedwith reference to FIGS. 37 to 40 may be formed in the silicon interposerto configure the digital filter 1002P as described above. Although theexample of using the well region WEL as the electrode in the MOScapacitive element (equivalently, MOS diode element) has been described,the invention is not limited to the well region WEL but the siliconsubstrate SSB may be used. Further, the semiconductor region DFRconfiguring the PN junction diode element may be formed in, not the wellregion, but in the silicon substrate SSB.

Tenth Embodiment

FIG. 41 is a block diagram illustrating the configuration of asemiconductor device according to a tenth embodiment. In the tenthembodiment, as described in the ninth embodiment, the case that thedigital filter 1002P is formed in the logic semiconductor chip LCH-1will be described.

FIG. 41 is similar to FIG. 37 related to the ninth embodiment. First,the part of the same configuration of FIG. 41 as that of FIG. 37 will bedescribed. The silicon substrate SSB, the well region WEL, the signallines SSP-L30 and SSN-L30, the voltage wire SSG-L30, the microbumpelectrodes LCH-PDG, LCH-PDN, and LCH-PDP, the contact CT1, and the bumpSMB in FIG. 41 are the same as those in FIG. 37. Consequently, theirdescription will not be repeated.

In FIG. 37, above the well region WEL, the delay line SSD-L10 isdisposed via the insulating film GIO, and the ends of the delay lineSSD-L10 are coupled to the signal lines SSN-L30 and SSP-L30. On theother hand, in the tenth embodiment, a plurality of delay lines aredisposed via the insulating film GIO above the well region WEL. Desirednumber of delay lines are selected from the plurality of delay lines,one end of each of the selected delay lines is coupled to the signalline SSP-L30, and the other end of the selected delay line is coupled tothe signal line SSN-L30. Consequently, the selected number of delaylines are coupled in parallel between the signal lines SSP-L30 andSSN-L30. As a result, the digital filter 1002P having the characteristicof an arbitrary inverse transfer function can be provided.

FIG. 41 illustrates the case where the number of delay lines disposedabove the well region via the insulating film GIO is four. In thediagram, reference numerals SSD10-L10 to SSD13-L10 are designated to thefour delay lines. Ends of each of the delay lines SSD10-L10 to SSD13-L10are coupled to the signal lines SSN-L30 and SSP-L30 via the contacts CT1via a switch array SAR1.

The switch array SAR1 has MOSFETs S10A to S13A and MOSFETs S10B to S13B.The MOSFETs S10A to S13A and the MOSFETs S10B to S13B are paired and thenumber of pairs corresponding to the number of delay lines are provided.Specifically, the MOSFETs S10A and S10B are paired and the paircorresponds to the delay line SSD10-L10. The MOSFETs S11A and S11B arepaired and the pair corresponds to the delay line SSD11-L10. Similarly,the MOSFETs S12A and S12B are paired and the pair corresponds to thedelay line SSD12-L10. The MOSFETs S13A and S13B are paired and the paircorresponds to the delay line SSD13-L10.

One end of the delay line SSD10-L10 is coupled to the signal lineSSP-L30 via the MOSFET S10A in the corresponding pair, and the other endis coupled to the signal line SSN-L30 via the MOSFET S10B in thecorresponding pair. One end of the delay line SSD11-L10 is coupled tothe signal line SSP-L30 via the MOSFET S11A in the corresponding pair,and the other end is coupled to the signal line SSN-L30 via the MOSFETS11B in the corresponding pair. Similarly, one end of the delay lineSSD12-L10 is coupled to the signal line SSP-L30 via the MOSFET S12A inthe corresponding pair, and the other end is coupled to the signal lineSSN-L30 via the MOSFET S12B in the corresponding pair. Further, one endof the delay line SSD13-L10 is coupled to the signal line SSP-L30 viathe MOSFET S13A in the corresponding pair, and the other end is coupledto the signal line SSN-L30 via the MOSFET S13B in the correspondingpair.

The MOSFETs configuring the switch array SAR1 are set to the on state inaccordance with delay line selection information stored in a delay lineselection register SREG1. For example, in the case where the delay lineselection information designates the delay line SSD10-L10, by selectionsignals SELA1 and SELB1 (each made of four bits) from the delay lineselection register SREG1, the MOSFETs S10A and S10B configuring the paircorresponding to the delay line SSF-L10 are turned on, and the remainingMOSFETs S11A to S13A and S11B to S13B are turned off. In the case wherethe delay line selection information designates the delay linesSSD10-L10 and SSD12-L10, by the selection signals SELA1 and SELB1 fromthe delay line selection register SREG1, the MOSFETs S10A, S10B, S12Aand S12B configuring the pairs corresponding to the delay lines areturned on, and the remaining MOSFETs S11A, S11B, S13A, and S13B areturned off. In such a manner, one or more of an arbitrary number ofpairs of MOSFETs are turned on by the delay line selection informationstored in the delay line selection register SREG1.

For example, when only the MOSFETs S10A and S10B are turned on, one endof the delay line SSD10-L10 is coupled to the signal line SSP-L30, andthe other end of the delay line SSD10-L10 is coupled to the signal lineSSN-L30. At this time, when the MOSFETs S12A and S12B are also turnedon, one end of each of the delay lines SSD10-L10 and SSD12-L10 iscoupled to the signal line SSP-L30, and the other end of each of thedelay lines SSD10-L10 and SSD12-L10 is coupled to the signal lineSSN-L30. In such a manner, one or more arbitrary delay lines areselected, one end of the one delay line or each of the plural delaylines is coupled to the signal line SSP-L30, and the other end iscoupled to the signal line SSN-L30.

The delay lines SSD10-L10 to SSD13-L10 having a desired delay amount anda desired signal loss are provided. Delay line selection information ofselecting one or more delay lines from the delay lines SSD10-L10 toSSD13-L10 is obtained in accordance with a proper loss amount of thedelay element DLN, and the obtained delay line selection information isstored in a delay line selection register SREG1. Consequently, one orplural delay lines designated by the delay line selection informationstored in the delay line selection register SREG1 is/are selected fromthe delay lines SSD10-L10 to SSD13-L10 and coupled between the signallines SSP-L30 and SSN-L30.

The configuration illustrated in FIG. 41 is suitable to the case wherethe bit rate of differential signals propagating through the signal lineis almost fixed. In this case, switching of delay lines by the delayline selection signal is used, for example, to adjust the loss amount ofthe delay element DLN. At this time, as will be described later in amodification, the delay amount of the delay element DLN is finelyadjusted by adjusting bias information.

Since one or plural delay lines selected functions as the delay elementDLN, in a case such that strength of equalization required changes, theinverse transfer function of the digital filter 1002P can be changeddynamically. Even at the same bit rate, for example, when the length ofthe signal line changes, attenuation which occurs in a signal channelbetween transmission and reception changes. When the attenuationincreases, strong equalization operation is necessary. When theattenuation decreases, weak equalization operation is necessary. To makethe equalization operation stronger, it is sufficient to select a delayline so that the loss amount of the delay element DLN decreases. On theother hand, to weaken the equalization operation, it is sufficient toselect a delay line so that the loss amount of the delay element DLNincreases.

Further, in the tenth embodiment, the voltage supplied to the wellregion WEL can be arbitrarily changed. Specifically, in the logicsemiconductor chip LCH-1, in addition to the delay line selectionregister SREG1 and the switch array SAR1, a power supply circuit RG anda bias voltage register BREG are provided. On the basis of biasinformation stored in the bias voltage register BREG, the power supplycircuit RG supplies, for example, as the bias voltage, a voltage betweenthe predetermined voltage Vs and a voltage Vd whose voltage value isdifferent from the predetermined voltage Vs to the well region WEL. Bythe operation, the voltage of the well region WEL can be set to anarbitrary voltage value. In the case of the equivalent circuitillustrated in FIG. 27B, the values of the capacitance C and theconductance G which are coupled in parallel to each other are changed bychanging the bias voltage of the well region WEL. For example, in thecase of equivalently regarding as a MOS diode element, by changing thebias voltage for inverse biasing the MOS diode element, the values ofthe capacitance C and the conductance G can be controlled. Therefore,the characteristic of the inverse transfer function of the delay elementDLN can be changed.

In the tenth embodiment, by delay line selection information, theresistance R of the equivalent circuit illustrated in FIG. 27B is mainlychanged. By the bias information, the capacitance C and the conductanceG o the equivalent circuit are mainly changed. Consequently, the inversetransfer function can be adjusted with higher precision, andequalization can be performed with high precision.

Although the case of adjusting the delay element DLN by using both thedelay line selection information and the bias information has beendescribed in the tenth embodiment, the invention is not limited to thecase. The delay element DLN may be adjusted by one of the delayselection information and the bias information. In the case ofperforming adjustment by the bias information, it is arranged not tosupply the predetermined voltage Vs to the voltage wire SSG-L30 orarranged to electrically separate the voltage wire SSG-L30 and the wellregion WEL from each other.

Modification

FIG. 42 is a block diagram illustrating the configuration of asemiconductor device according to a modification of the tenthembodiment. Since FIG. 42 is similar to FIG. 41, only the differentpoints will be mainly described. The parts different from FIG. 41 are aswitch array, a delay line, and a delay line selection register. Sincethe other part is the same as that of FIG. 41, its description will notbe repeated in principle.

The configuration illustrated in FIG. 41 is suitable to the case wherethe bit rate of differential signals is almost fixed. In this case,since the bit rate is almost fixed, switching of delay lines by a delayline selection signal is used to adjust, for example, the loss amount ofthe delay element DLN. The bias information is used to finely adjust thedelay amount of the delay element DLN.

On the other hand, the modification is suitable to the case where theyare a plurality of bit rates of differential signals, and the bit ratechanges dynamically.

Also in the embodiment, like FIG. 41, delay lines SSD20-L10 to SSD23-L10are disposed above the well region WEL via the insulating film GIO. Eachof the delay lines SSD20-L10 to SSD23-L10 is formed so as to have anappropriate delay amount and an appropriate loss amount in advance sothat it operates the delay element DLN suited to each of bit rates. Forexample, the delay line SSD20-L10 is formed so that the delay lineSSD20-L10 is suitable as the delay element DLN at a first bit rate. Thedelay line SSD21-L10 is formed so that the delay line SSD21-L10 issuitable as the delay element DLN at a second bit rate which isdifferent from the first bit rate. Similarly, the delay line SSD22-L10is formed so as to be adapted to a third bit rate, and the delay lineSSD23-L10 is formed so as to be adapted to a fourth bit rate.

A switch array SAR2 has, like the switch array SAR1, a set of MOSFETscorresponding to the delay lines SSD20-L10 to SSD23-L10. In FIG. 42, aset corresponding to the delay line SSD20-L10 is comprised of MOSFETsS20A and S20B, and a set corresponding to the delay line SSD21-L10 iscomprised of MOSFETs S21A and S21B. Similarly, a set corresponding tothe delay line SSD22-L10 is comprised of MOSFETs S22A and S22B, and aset corresponding to the delay line SSD23-L10 is comprised of MOSFETsS23A and S23B.

One end of each of the delay lines SSD20-L10 to SSD23-L10 is coupled tothe signal line SSP-L30 via the MOSFETs S20A to S23A of thecorresponding set. On the other hand, the other end of each of the delaylines SSD20-L10 to SSD23-L10 is coupled to the signal line SSN-L30 viathe MOSFETs S20B to S23B of the corresponding set.

In the modification, a delay line selection register SREG2 stores delayline selection information for selecting one of the delay linesSSD20-L10 to SSD23-L10. By selection signals SELA2 and SELB2 (each madeof four bits) based on the delay line selection information, a delayline designated by the delay line selection information is selected, andthe selected delay line is coupled between the signal lines SSP-L30 andSSN-L30 by the switch array SAR2.

For example, when the delay line SSD20-L10 is designated by the delayline selection information, the MOSFETs S20A and S20B configuring theset corresponding to the selection line SSD20-L10 are turned on by theselection signals SELA2 and SELB2, and the remaining MOSFETs S21A toS23A and S21B to S23B are turned off. When the delay line SSD22-L10 isdesignated by the delay line selection information, the MOSFETs S22A andS22B configuring the set corresponding to the selection line SSD22-L10are turned on by the selection signals SELA2 and SELB2, and theremaining MOSFETs S20A, S21A, S23A, S20B, S21B, and S23B are turned off.In such a manner, only MOSFETs configuring a set are turned on.

By turning on the MOSFETs of the set corresponding to the delay linedesignated by the delay line selection information, only one delay linedesignated by the delay line selection signals in the delay linesSSD20-L10 to SSD23-L10 is electrically coupled between the signal linesSSN-L30 and SSP-L30.

Consequently, even the bit rate of the differential signal changes, forexample, from a first bit rate to a third bit rate, by changing thedelay line selection information stored in the delay line selectionregister SREG2 from the information designating the delay line SSD20-L10corresponding to the first bit to the information designating the delayline SSD23-L10 corresponding to the third bit rate, equalization can beproperly performed even when the bit rate changes.

As described in FIG. 41, by changing the bias information stored in thebias voltage register BREG, the delay amount of the delay element DLNcan be finely adjusted at each bit rate. Also in the modification, inthe case of performing the adjustment with the bias information, it isarranged not to supply the predetermined voltage Vs to the voltage wireSSG-L30 or arranged to electrically separate the voltage wire SSG-L30and the well region WEL.

Although the example of configuring the delay element DLN by using thedelay line disposed in the well region WEL via the insulating film GIOhas been described in the tenth embodiment and its modification, thepresent invention is not limited to the example. For example, in thetenth embodiment and its modification, a delay line which isohmic-coupled to the semiconductor region DEF formed in the well regionWEL may be used as a delay line as described in the modification of theninth embodiment.

From the viewpoint of controlling the delay element DLN, it can beregarded that the control circuit is comprised of the delay lineselection register SREG1, the bias voltage register BREG, the powersupply circuit RG, and the switch array SAR1 illustrated in FIG. 41.Similarly, it can be regarded that the control circuit is comprised alsoof the delay line selection register SREG2, the bias voltage registerBREG, the power supply circuit RG, and the switch array SAR2 illustratedin FIG. 42.

In FIGS. 41 and 42, the delay element DLN has a plurality of delaylines. Since the time of signal delay in the delay element DLN, that is,a round-trip signal delay can be changed by the control circuit, thedelay element DLN can be regarded as a variable delay element. In otherwords, it can be regarded that the delay time of the delay element DLNas a variable delay element is determined by the control circuit. Inthis case, by determining bias voltage supplied to the diode elementincluded in the delay element DLN and/or a delay line coupled betweenthe signal lines by the control circuit, the delay time of the variabledelay element is determined.

Although the delay element DLN to which a pair of differential signalsis supplied has been described as an example in the tenth embodiment,the delay line selection register, the bias voltage register, the powersupply circuit, and the switch array described in the tenth embodimentcan be also applied to the third or fourth embodiment. That is, they canbe also applied to a delay element corresponding to a single-end signal.In this case, in the third or fourth embodiment, a plurality of delaylines are provided and a delay element is comprised of a delay linedesignated by the delay line selection information stored in the delayline selection register. In a manner similar to the tenth embodiment, awell region is formed in the silicon substrate SSB and, by adjusting thevoltage of the well region by the power supply circuit and the biasvoltage register, the delay amount of the delay element is adjusted.

The delay element DLN described in the first to sixth embodiments iscoupled to a signal line. From the viewpoint of coupling to a signalline, the delay element can be considered as a kind of so-called shortstubs. However, the delay element described in the embodiments is quitedifferent from a short stub for the following reason.

A short stub does not function sufficiently when its loss is large. Onthe other hand, the delay element is set so that its loss becomes large.In the case of the equivalent circuit illustrated in FIG. 1B, the delayelement is set so that the resistance R or the parallel conductance Gbecomes large. In principle, the length of a short stub is set to around¼ of an electromagnetic wave length to the input signal frequency. Onthe other hand, the delay element is not directly related to theelectromagnetic wave length and its length is not determined by theelectromagnetic wave length. The length of the delay element isdetermined by, for example, time of a round-trip signal delay. Theround-trip signal delay is also determined by a fraction of an integerof the one data width interval, not by the electromagnetic wave length.

Further, when a short stub functions as a short stub, the line lengthfrom a signal source to the short stub which is long enough to beregarded as a transmission path is necessary. For example, the linelength from the signal source to the short stub has to be set to ¼ ofthe electromagnetic wave length. On the other hand, desirably, the delayelement is coupled near the transmission buffer circuit (signal source)or the reception buffer circuit so that it is not regarded as atransmission path. That is, it is desirable to couple the delay elementin a position where the delay element does not function as a short stub.

In the first to sixth embodiments, the example of forming the digitalfilter in the interposer or the semiconductor chip has been described.It is also possible to form the digital filter 1002 described in thefirst to sixth embodiments in a small-sized interposer and bury thesmall-sized interposer in the package substrate or the print substrate.

Although the example of wired-OR coupling the delay element DLN near apair of input terminals of the reception buffer circuit has beendescribed in the seventh to tenth embodiments, the present invention isnot limited to the example. For example, one end of the delay elementDLN may be wired-OR coupled to one of a pair of output terminals of thetransmission buffer circuit, and the other end of the delay element DLNmay be wired-OR coupled to the other output terminal of the transmissionbuffer circuit. In this case, before the components of differentialsignals are transmitted by a pair of signal lines, waveform shaping withthe inverse transfer function to equalize transfer functions of the pairof signal lines is performed. Therefore, the waveform of thedifferential signal components with reduced distortion is supplied to apair of input terminals of the reception buffer circuit. At this time,the components of the common mode signal are transmitted to a pair ofinput terminals of the reception buffer circuit without being equalized.As a result, the components of the common mode signal can be preventedfrom being erroneously recognized.

In the seventh and eighth embodiments, the example of using the wiringpattern formed in the interposer as the delay line has been described.However, the invention is not limited to the example. For example, theinterposer INS-1 is not limited to a silicon interposer but may be aninterposer using an organic substrate or glass substrate. The wiringpattern formed in the logic semiconductor chip may be used as a delayline. Further, a small-sized semiconductor chip (semiconductor chip forequalization) in which the delay line described in the seventh to tenthembodiments and the voltage wire disposed along the delay line isprovided, and the semiconductor chip for equalization may be buried inthe package substrate PPS-1 and/or the interposer INS-1. That is, thesemiconductor chip for equalization may be disposed so as to be buriedbetween the first main surface PPF1 and the second main surface PPF2 ofthe package substrate PPS-1. The semiconductor chip for equalization maybe disposed so as to be buried between the first main surface INF1 andthe second main surface of the interposer INS-1.

Supplemental Notes

In the specification, a plurality of inventions are disclosed. Some ofthem are described in the scope of claims but other inventions are alsodisclosed. Some representative ones will be described as follows.

(A) A semiconductor device comprises:

a pair of differential signal lines;

a first circuit which is coupled to an end of each of the pair of thedifferential signal lines and to/from which differential signals aresupplied from/to the pair of differential signal lines; and

a delay element having one end which is wired-OR coupled to an end ofone of the pair of differential signal lines and another end which iswired-OR coupled to an end of the other differential signal line of thepair of differential signal lines, and which shapes waveform of adifferential signal at the end of the pair of differential signal lines.

(B) In the semiconductor device described in (A), the delay element hasa delay line having a pair of ends, one end of the delay line is, as oneend of the delay element, is wired-OR coupled to an end of one of thedifferential signal lines, and the other end of the delay line is, asthe other end of the delay element, is wired-OR coupled to an end of theother differential signal line, and

the semiconductor device has a voltage wire which is disposed along thedelay line and to which predetermined voltage is supplied.

(C) In the semiconductor device described in (B),

the delay line is set so that a round-trip signal delay between a signalinput to the one end or the other end and an output signal output fromthe one end or the other end becomes twice or a fraction of an integerof time of one data with interval of the signal.

(D) The semiconductor device described in (A), further comprises a diodeelement, in which the delay element is comprised of the diode element.(E) In the semiconductor device described in (A), the delay element is avariable delay element in which delay time can be changed, and

the semiconductor device includes a control circuit determining delaytime of the variable delay element.

(F) In the semiconductor device described in (E), the semiconductordevice has a diode element, the variable delay element includes thediode element, and bias voltage supplied to the diode element is set bythe control circuit.(G) In the semiconductor device described in (E), the variable delayelement has a plurality of delay lines, and a delay line selected by thecontrol circuit is coupled between ends of the pair of differentialsignal lines.(H) A semiconductor device comprises:

a first semiconductor chip having a main surface over which a pair ofelectrodes to/from which differential signals are input/output; and

an interposer having a first main surface over which a pair ofelectrodes is formed and a second main surface which is opposed to thefirst main surface and over which a pair of second electrodeselectrically coupled to the pair of the first electrodes is formed, andmounted so that a main surface of the first semiconductor chip faces thefirst main surface so that the pair of electrodes of the firstsemiconductor chip are coupled to the pair of first electrodes,

in which when differential signals are transmitted between the pair ofsecond electrodes and the pair of electrodes, the differential signalsare shaped by a delay line having one end which is wired-OR coupled toone of the pair of electrodes and the other end which is wired-ORcoupled to the other electrode of the pair of electrodes, thedifferential signals are shaped.

(I) The semiconductor device described in (H), further comprises avoltage wire which is disposed along the delay line and to whichpredetermined voltage is supplied.(J) In the semiconductor device described in (I), the delay line and thevoltage wire are wires formed in the first semiconductor chip.(K) In the semiconductor device described in (I), the delay line and thevoltage wire are wires formed in the interposer.(L) The semiconductor device described in (I), further comprises asemiconductor chip for equalization in which the delay line and thevoltage wire are formed.(M) In the semiconductor device described in (L), the semiconductor chipfor equalization is disposed between the first and second main surfacesof the interposer.(N) A semiconductor device comprises:

a first semiconductor chip having a main surface over which a pair ofelectrodes to/from which differential signals are input/output isformed;

a first interposer having a first main surface over which a pair offirst electrodes is formed, a second main surface which faces the firstmain surface and over which a pair of second electrodes electricallycoupled to the pair of first electrodes is formed, and mounted so thatthe main surface of the first semiconductor chip faces the first mainsurface so that the pair of electrodes of the first semiconductor chipis coupled to the first electrode;

a substrate having a main surface opposed to the second main surface ofthe first interposer, a pair of third electrodes formed over the mainsurface, a pair of fourth electrodes formed over the main surface, and awiring pattern electrically coupling the pair of third electrodes andthe pair of fourth electrodes; and

a delay line having one end coupled to one of the pair of electrodes ofthe first semiconductor chip and another end coupled to the otherelectrode of the pair of electrodes of the first semiconductor chip,

in which the pair of third electrodes is electrically coupled to thepair of second electrodes and, when a signal is transmitted between theset of fourth electrodes and the pair of electrodes of the firstsemiconductor chip, the differential signal is shaped by the delay line.

(O) The semiconductor device described in (N), further comprises:

a second semiconductor chip having a main surface over which a pair ofelectrodes is formed; and

a second interposer having a first main surface over which a pair offifth electrodes is formed and a second main surface which is opposed tothe first main surface and over which a pair of sixth electrodeselectrically coupled to the pair of fifth electrodes is formed,

the main surface of the second semiconductor chip being mounted so as tobe opposed to the first main surface so that the pair of electrodes ofthe second semiconductor chip is electrically coupled to the pair offifth electrodes,

in which a second main surface of the second interposer faces the mainsurface of the substrate, the pair of sixth electrodes of the secondinterposer is electrically coupled to the pair of fourth electrodes,

the first semiconductor chip has a first circuit amplifying a signalfrom the pair of electrodes of the first semiconductor chip, and thesecond semiconductor chip has a second circuit outputting a serialsignal to the pair of electrodes of the second semiconductor chip.

(P) The semiconductor device described in (O), further comprises avoltage wire which is disposed along the delay line and to whichpredetermined voltage is supplied.

Although the invention achieved by the inventors of the presentinvention has been concretely described on the basis of the embodiments,obviously, the present invention is not limited to the embodiments butcan be variously changed without departing from the gist. For example, alogic semiconductor chip was described as a semiconductor chip. However,the semiconductor chip is not limited to a logic semiconductor chip. Inaddition to the digital filter described in the first to sixthembodiments, a semiconductor chip may be provided with an analog filtercircuit and/or a digital filter circuit. In this case, by the analogfilter circuit and/or the digital filter circuit, precision ofrestoration of signals can be further improved. For example, the eyepatterns illustrated in FIGS. 9 and 10 can be made visible.

What is claimed is:
 1. A semiconductor device comprising: a signal linehaving one end and another end; a first circuit coupled to the one endof the signal line, and to which a signal is to be supplied from theanother end of the signal line; and a delay element wired-OR coupled tothe one end of the signal line, and shaping a waveform of the signalsuch that a tail part of the waveform of the signal at the one end ofthe signal line is eliminated.
 2. The semiconductor device according toclaim 1, wherein the delay element includes a delay line having: one endwired-OR coupled to the one end of the signal line, and another endcoupled to a predetermined voltage, and wherein the waveform of thesignal at the one end of the signal line is shaped by a reflected signalcorresponding to a signal input to the one end of the delay line.
 3. Thesemiconductor device according to claim 2, wherein the delay line iscomprised such that a signal delay between the signal input to the oneend of the delay line and the reflected signal output from the one endof the delay line is a specified fraction of an integer of one datawidth interval.
 4. The semiconductor device according to claim 3 furthercomprising: a second circuit coupled to the another end of the signalline, and supplying the signal to the first circuit.
 5. Thesemiconductor device according to claim 4, wherein the delay line iscomprised of an inductance, a resistance, a capacitance, and aconductance.
 6. The semiconductor device according to claim 1, whereinthe signal line has a first signal line and a second signal linetransmitting a differential signal which complementarily changes witheach other, wherein the first circuit has a differential circuit coupledto each of one end of the first signal line and one end of the secondsignal line, and wherein the delay element has: one end wired-OR coupledto the one end of the first signal line, and the other end wired-ORcoupled to the one end of the second signal line.